Wide dynamic range and high speed voltage mode sensing for a multilevel digital non-volatile memory

ABSTRACT

A high speed voltage mode sensing is provided for a digital multibit non-volatile memory integrated system. An embodiment has a local source follower stage followed by a high speed common source stage. Another embodiment has a local source follower stage followed by a high speed source follower stage. Another embodiment has a common source stage followed by a source follower. An auto zeroing scheme is used. A capacitor sensing scheme is used. Multilevel parallel operation is described.

CROSS-REFERENCE TO RELATED APPLICATION

This is a continuation-in-part of application Ser. No. 09/929,542, filedAug. 13, 2001, which is a division of application Ser. No. 09/231,928filed Jan. 14, 1999, issued as U.S. Pat. No. 6,282,145, the subjectmatter of each of these applications is incorporated herein byreference.

FIELD OF THE INVENTION

This invention relates in general to semiconductor memories, and, inparticular, to the design and operation of multilevel nonvolatilesemiconductor memories.

BACKGROUND OF THE INVENTION

As the information technology progresses, the demand for high densitygiga bit and tera bit memory integrated circuits is insatiable inemerging applications such as data storage for photo quality digitalfilm in multi-mega pixel digital camera, CD quality audio storage inaudio silicon recorder, portable data storage for instrumentation andportable personal computers, and voice, data, and video storage forwireless and wired phones and other personal communicating assistants.

The nonvolatile memory technology such as ROM (Read Only Memory), EEPROM(Electrical Erasable Programmable Read Only Memory), or FLASH is often atechnology of choice for these application due to its nonvolatilenature, meaning it still retains the data even if the power supplied toit is removed. This is in contrast with the volatile memory technology,such as DRAM (Dynamic Random Access Memory), which loses data if thepower supplied to it is removed. This nonvolatile feature is very usefulin saving the power from portable supplies, such as batteries. Untilbattery technology advances drastically to ensure typical electronicsystems to function for a typical operating lifetime, e.g., 10 years,the nonvolatile technology will fill the needs for most portableapplications.

The FLASH technology, due to its smallest cell size, is the highestdensity nonvolatile memory system currently available. The advance ofthe memory density is made possible by rapidly advancing the processtechnology into the realm of nano meter scale and possibly into theatomic scale and electron scale into the next century. At the presentsub-micro meter scale, the other method that makes the superhigh-density memory system possible is through the exploitation of theanalog nature of a storage element.

The analog nature of a flash or nonvolatile storage element provides, bytheory, an enormous capability to store information. For example, if oneelectron could represent one bit of information then, for one typicalconventional digital memory cell, the amount of information is equal tothe number of electrons stored, or approximately a few hundredthousands. Advances in device physics exploring the quantum mechanicalnature of the electronic structure will multiply the analog informationmanifested in the quantum information of a single electron even further.

The storage information in a storage element is hereby defined as adiscrete number of storage levels for binary digital signal processingwith the number of storage levels equal to 2^(N) with N equal to thenumber of digital binary bits. The optimum practical number of discretelevels stored in a nonvolatile storage element depends on the innovativecircuit design method and apparatus, the intrinsic and extrinsicbehavior of the storage element, all within constraints of a definiteperformance target, such as product speed and operating lifetime, with acertain cost penalty.

At the current state of the art, all the multilevel systems are onlysuitable for medium density, i.e. less than a few tens of mega bits, andonly suitable for a small number of storage levels per cell, i.e., lessthan four levels or two digital bits.

As can be seen, memories having high storage capacity and fast operatingspeed are highly desirable.

SUMMARY OF THE INVENTION

This invention describes the design method and apparatus for a superhigh density nonvolatile memory system capable of giga to tera bits asapplied to the array architecture, reference system, and decodingschemes to realize the optimum possible number of storage levels withinspecified performance constraints. Method and apparatus for multilevelprogram and sensing algorithm and system applied to flash memory is alsodescribed in this invention. Details of the invention and alternativeembodiments will be made apparent by the following descriptions.

The invention provides array architectures and operating methodssuitable for a super high density, in the giga to tera bits, formultilevel nonvolatile “green” memory integrated circuit system. “Green”refers to a system working in an efficient and low power consumptionmanner. The invention solves the issues associated with super highdensity multilevel memory system, such as, precision voltage control inthe array, severe capacitive loading from MOS transistor gates andparasitics, high leakage current due to memory cells and from cells tocells, excessive power consumption due to large number of gates andparasitics, and excessive memory cell disturbances due to large memorydensity.

An aspect of the invention provides an Inhibit and Select SegmentationScheme that makes use of a truly-floating-bitline scheme to greatlyreduce the capacitance from junctions and parasitic interconnects to asmall value.

The invention also provides a Multilevel Memory Decoding scheme which iscapable of greater than 10-bit multilevel operation. The MultilevelMemory Decoding Scheme includes the Power Supply Decoded DecodingScheme, the Feedthrough-to-Memory Decoding Scheme, and theFeedthrough-to-Driver Decoding Scheme. The Multilevel Memory Decodingscheme also includes a “winner-take-all” Kelvin Decoding Scheme, whichprovides precise bias levels for the memory at a minimum cost. Theinvention also provides a constant-total-current-program scheme. Theinvention also provides fast-slow and 2-step ramp rate controlprogramming. The invention also presents reference system method andapparatus, which includes the Positional Linear Reference System,Positional Geometric Reference System, and the Geometric CompensationReference System. The invention also describes apparatus and method ofmultilevel programming, reading, and margining.

A sense amplifier system includes local sense amplifiers coupled tomemory subarrays and global sense amplifiers coupled to groups of localsense amplifiers.

Method and apparatus described herein are applicable to digitalmultilevel as well as analog multilevel system.

The foregoing, together with other aspects of this invention, willbecome more apparent when referring to the following specification,claims, and accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a cross section of a source side injection flash memory cell.

FIG. 1B is a transistor symbol corresponding to the source sideinjection flash memory cell shown in FIG. 1A.

FIG. 1C is a block diagram of a nonvolatile multilevel memory system.

FIG. 1D is a block diagram of an electronic camera system utilizing anonvolatile multilevel memory system.

FIG. 1E is a block diagram of an electronic audio system utilizing anonvolatile multilevel memory system.

FIG. 2A is a block diagram of super high-density nonvolatile multilevelmemory integrated circuit system.

FIG. 2B is a block diagram of flash power management unit.

FIG. 2C shows voltage mode sensing.

FIG. 3A is a block diagram of super high-density nonvolatile multilevelarray architecture.

FIG. 3B is a page select circuit, which together with the segment selectdecoder selects one bitline at a time for each y-driver.

FIG. 3C is a block diagram of a multilevel sub-array block.

FIG. 4A is one embodiment of a nonvolatile multilevel array unit ofinhibit and select segmentation.

FIG. 4B shows an alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4C shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4D shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4E shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 4F shows another alternate embodiment of the inhibit and selectsegmentation scheme.

FIG. 5A is a cross section of inhibit and select segmentationinterconnection.

FIG. 5B is a cross section of another embodiment of inhibit and selectsegmentation interconnection.

FIG. 5C is a 2-step ramp rate control and fast-slow ramp rate control.

FIG. 6 shows a block diagram of multilevel decoding.

FIG. 7 shows one segment decoder that includes segmented power supplydecoder, segmented bitline select decoder, inhibit decoder, segmentedpredecoded common line decoder, and control gate and control linedecoder.

FIG. 8 shows a segmented power supply decoder.

FIG. 9A shows a segmented bitline decoder.

FIG. 9B shows a segmented inhibit decoder.

FIG. 9C shows a segmented predecoded common line decoder.

FIG. 10 shows a sub-block decoder for control gate and common linemultilevel decoder.

FIG. 11A shows a sub-block of the circuit in FIG. 10 for four controlgates and one common line multilevel decoder.

FIG. 11B shows another embodiment of sub-block for four control gatesand one common line multilevel decoder with winner-take-all Kelvinconnection.

FIG. 11C shows a circuit for one common line driver.

FIG. 12 shows a scheme of the feedthrough-to-driver andfeedthrough-to-memory multilevel precision decoding.

FIG. 13 shows a block diagram of a multilevel reference system.

FIG. 14 shows details of a block diagram of a multilevel referencesystem.

FIG. 15 shows a reference detection scheme.

FIG. 16 shows positional linear reference system.

FIG. 17 shows a positional geometric reference system.

FIG. 18 shows an embodiment of geometric compensation reference scheme.

FIG. 19A shows voltage levels for program verify, margin, read, andrestore for one embodiment of the current invention.

FIG. 19B shows voltage levels for program verify, margin, read, andrestore for an alternative embodiment of the current invention.

FIG. 20 shows an embodiment of flow diagram of the page programmingcycle.

FIG. 21 shows an embodiment of flow diagram after page programmingbegins.

FIG. 22A shows a continuation of flow diagram after page programmingbegins.

FIG. 22B shows an alternative embodiment of continuation of flow diagramafter page programming begins shown in FIG. 22A.

FIG. 22C shows an alternate embodiment of the flow diagram shown in FIG.22B.

FIG. 23 shows an embodiment of flow diagram of the page read cycle.

FIG. 24 shows a continuation of flow diagram of the page read cycle inFIG. 23.

FIG. 25 shows a continuation of flow diagram of the page read cycle inFIG. 24.

FIG. 26 shows details of an embodiment of a single y-driver YDRVS 110S.

FIG. 27 shows details of a latch block, a program/read control block,and program/program inhibit block included in the single y-driver YDRVS110S.

FIG. 28 is a block diagram illustrating a memory system for a multilevelmemory;

FIG. 29A is a block diagram illustrating an inverter mode sensingcircuit.

FIG. 29B is a block diagram illustrating a voltage mode sensing circuit.

FIG. 30 is a block diagram illustrating a wide range, high speed voltagemode sensing circuit.

FIG. 31 is a block diagram illustrating a wide range, high speed modesensing circuit having a local source follower stage and a global commonsource stage.

FIG. 32 is a block diagram illustrating a wide range, high speed modesensing circuit with a local PMOS source follower stage and a globalsource follower stage.

FIG. 33 is a block diagram illustrating a wide range, high speed modesensing circuit with a local NMOS source follower stage and a globalsource following stage.

FIG. 34 is a block diagram illustrating a global sense amplifier havingan auto zeroing function.

FIG. 35 is a block diagram illustrating an auto zero sense amplifier.

DESCRIPTION OF THE SPECIFIC EMBODIMENTS

Memory Cell Technology

To facilitate the understanding of the invention, a brief description ofa memory cell technology is described below. In an embodiment theinvention applies to Source Side Injection (SSI) flash memory celltechnology, which will be referred to as SSI flash memory celltechnology. The invention is equally applicable to other technologiessuch as drain-side channel hot electron (CHE) programming (ETOX),P-channel hot electron programming, other hot electron programmingschemes, Fowler-Nordheim (FN) tunneling, ferro-electric memory, andother types of memory technology.

A cell structure of one typical SSI flash cell is symbolically shown inFIG. 1A. Its corresponding transistor symbol is shown in FIG. 1B. Thecell is made of two polysilicon gates (abbreviated as poly), a floatinggate poly FG 100F and a control gate poly CG 100C. The control gate CG100C also acts as a select gate that individually select each memorycell. This has the advantage of avoiding the over erase problem which istypical of stacked gate CHE flash cell. The floating gate has a poly tipstructure that points to the CG 100C, this is to enhance the electricfield from the FG 100F to the CG 100C which allows a much lower voltagein FN erase without using a thin interpoly oxide.

The thicker interpoly oxide leads to a higher reliability memory cell.The cell is also fabricated such that a major portion of the FG 100Foverlaps the source junction 100S. This is to make a very high couplingratio from the source 100S to FG 100F, which allows a lower erasevoltage and is advantageous to the SSI programming, which will bedescribed shortly. A structural gap between the FG 100F and at CG 100Cis also advantageous for the efficient SSI programming.

The SSI flash memory cell enables low voltage and low power performancedue to its intrinsic device physics resulting from its device structure.The SSI flash cell uses efficient FN tunneling for erase and efficientSSI for programming. The SSI flash cell programming requires a smallcurrent in hundreds of nano amps and a moderate voltage range of ˜8 to11 volts. This is in contrast to that of a typical drain-side channelhot electron memory cell programming which requires current in hundredsof microamp to milliamp range and a voltage in the range of 11 to 13volts.

The SSI flash memory cell erases by utilizing Fowler-Nordheim tunnelingfrom the floating gate poly to the control gate poly by applying a higherase voltage on the control gate CG 100C, e.g., 8-13 volts, and a lowvoltage on the source 100S, e.g., 0-0.5 volts. The high erase voltagetogether with high coupling from the source to the floating gate createsa localized high electric field from the FG 100F tip to the CG 100C andcauses electrons to tunnel from the FG 100F to the CG 100C near the tipregion. The resulting effect causes a net positive charge on the FG100F.

The SSI flash memory cell programs by applying a high voltage on thesource 100S (herein also known as common line CL), e.g., 4-13 V, a lowvoltage on the CG 100C, e.g., 0.7-2.5 V, and a low voltage on the drain100D (herein also known as the bitline BL), e.g., 0-1V. The high voltageon the source 100S strongly couples to the FG to strongly turn on thechannel under the FG (it will be equivalently referred to as the FGchannel). This in turn couples the high voltage on the source 100Stoward the gap region. The voltage on the CG 100C turns on the channeldirectly under the CG 100C (it will be equivalently referred to as theCG channel). This in turn couples the voltage on the drain 100D towardthe gap region. Hence, the electrons flow from the drain junction 100Dthrough the CG channel, through the gap channel, through the FG channel,and finally arrive at the source junction.

Due to the gap structure between the CG 100C and the FG 100F, in thechannel under the gap, there exists a strong lateral electric field(EGAPLAT) 100G. As the EGAPLAT 100G reaches a critical field, electronsflowing across the gap channel become hot electrons. A portion of thesehot electrons gains enough energy to cross the interface between thesilicon and silicon dioxide into the silicon dioxide. And as thevertical field Ev is very favorable for electrons to move from thechannel to the FG 100F, many of these hot electrons are swept toward theFG 100F, thus, reducing the voltage on the FG 100F. The reduced voltageon the FG 100F reduces electrons flowing into the FG 100F as programmingproceeds.

Due to the coincidence of favorable Ev and high EGAPLAT 100G in the gapregion, the SSI memory cell programming is more efficient over that ofthe drain-side CHE programming, which only favors one field over theother. Programming efficiency is measured by how many electrons flowinto the floating gate as a portion of the current flowing in thechannel. High programming efficiency allows reduced power consumptionand parallel programming of multiple cells in a page mode operation.

Multilevel Memory Integrated Circuit System:

The challenges associated with putting together a billion transistors ona single chip without sacrificing performance or cost are tremendous.The challenges associated with designing consistent and reliablemultilevel performance for a billion transistors on a single chipwithout sacrificing performance or cost are significantly moredifficult. The approach taken here is based on the modularizationconcept. Basically everything begins with a manageable optimized basicunitary block. Putting appropriate optimized unitary blocks togethermakes the next bigger optimized block.

A super high density nonvolatile multilevel memory integrated circuitsystem herein described is used to achieve the performance targets ofread speed, write speed, and an operating lifetime with low cost. Readspeed refers to how fast data could be extracted from a multilevelmemory integrated circuit system and made available for external usesuch as for the system microcontroller 2001 shown in FIG. 1C which isdescribed later. Write speed refers to how fast external data could bewritten into a multilevel memory integrated circuit system. Operatinglifetime refers to how long a multilevel memory integrated circuitsystem could be used in the field reliably without losing data.

Speed is modularized based on the following concept, T=CV/I, whereswitching time T is proportional to capacitance C multiplied by thevoltage swing V divided by the operating current I. Methods andapparatuses are provided by the invention to optimize C, V, and I toachieve the required specifications of speed, power, and optimal cost toproduce a high performance high-density multilevel memory integratedcircuit system. The invention described herein makes the capacitanceindependent of memory integrated circuit density, to the first order,and uses the necessary operating voltages and currents in an optimalmanner.

A nonvolatile multilevel memory system is shown in FIG. 1C. A super highdensity nonvolatile multilevel memory integrated circuit (IC) system2000 is a digital multilevel nonvolatile flash memory integrated circuitcapable of storing 2^(N) storage levels per one memory cell, withN=number of digital bits. A system microcontroller 2001 is a typicalsystem controller used to control various system operations. Controlsignals (CONTROL SIGNALS) 196L, input/output bus (IO BUS) 194L, andready busy signal (R/BB) 196RB are for communication between the systemmicrocontroller 2001 and the super high density nonvolatile multilevelmemory integrated circuit system 2000.

An electronic camera system (SILICONCAM) 2008 utilizing super highdensity nonvolatile multilevel memory IC system 2000 is shown in FIG.1D. The system (SILICONCAM) 2008 includes an integrated circuit system(ECAM) 2005 and an optical lens block (LENS) 2004. The integratedcircuit system (ECAM) 2005 includes an image sensor (IMAGE SENSOR) 2003,an analog to digital converter block (A/D CONVERTER) 2002, a systemmicrocontroller 2001, and the multilevel memory IC system 2000. Theoptical lens block (LENS) 2004 is used to focus light into the IMAGESENSOR 2003, which converts light into an analog electrical signal. TheIMAGE SENSOR 2003 is a charge coupled device (CCD) or a CMOS sensor. Theblock (A/D CONVERTER) 2002 is used to digitize the analog electricalsignal into digital data. The microcontroller 2001 is used to controlvarious general functions such as system power up and down, exposuretime and auto focus. The microcontroller 2001 is also used to processimage algorithms such as noise reduction, white balance, imagesharpening, and image compression. The digital data is stored in themultilevel memory IC system 2000. The digital data can be down loaded toanother storage media through wired or wireless means. Future advancesin process and device technology can allow the optical block (LENS) 2004to be integrated in a single chip with the ECAM 2005.

An electronic audio system (SILICONCORDER) 2007 utilizing super highdensity nonvolatile multilevel memory IC system 2000 is shown in FIG.1E. The SILICONCORDER 2007 includes an integrated circuit system(SILICONAUDIO) 2006, a MICROPHONE 2012, and a SPEAKER 2013. The system(SILICONAUDIO) 2006 includes an anti-alias FILTER 2010, an A/D CONVERTER2002, a smoothing FILTER 2011, a D/A CONVERTER 2009, a systemmicrocontroller 2001, and the multilevel memory IC system 2000. TheFILTER 2010 and the FILTER 2011 can be combined into one filter block ifthe signals are multiplexed appropriately. The microcontroller 2001 isused to control various functions such as system power up and down,play, record, message management, audio data compression, and voicerecognition. In recording a sound wave, the MICROPHONE 2012 converts thesound wave into an analog electrical signal, which is filtered by theFILTER 2010 to reduce non-audio signals. The filtered analog signal isthen digitized by the A/D CONVERTER 2002 into digital data. The digitaldata is then stored in compressed or uncompressed form in the multilevelmemory IC system 2000. In playing back the stored audio signal, themicrocontroller 2001 first uncompresses the digital data if the data isin compressed form. The D/A CONVERTER 2009 then converts the digitaldata into an analog signal which is filtered by a smoothing filter(FILTER) 2011. The filtered output analog signal then goes to theSPEAKER 2013 to be converted into a sound wave. The signal filtering canbe done by digital filtering by the microcontroller 2001. Externaldigital data can be loaded into the multilevel memory IC system 2000through wired or wireless means. Future advances in process and devicetechnology can allow the MICROPHONE 2012 and the SPEAKER 2013 to beintegrated in a single chip with the SILICONAUDIO 2006.

A circuit block diagram of the super high density nonvolatile multilevelmemory integrated circuit system 2000 based on the concepts describedabove and also on ideas described below, is shown in FIG. 2A. For thepurpose of discussion, a giga bit nonvolatile multilevel memory chip isdescribed.

A circuit block 100 includes a regular memory array.

It includes a total of for example, 256 million nonvolatile memory cellsfor a 4-bit digital multilevel memory cell technology or 128 millioncells for a 8-bit digital multilevel memory cell technology. An N-bitdigital multilevel cell is defined as a memory cell capable of storing2^(N) levels. A reference array (MFLASHREF) 106 is used for thereference system. A redundancy array (MFLASHRED) 102 is used to increaseproduction yield by replacing bad portions of the regular memory arrayof the circuit block 100. An optional spare array (MFLASHSPARE) 104 canbe used for extra data overhead storage such as for error correction.

A y-driver block (YDRV) 110 including a plurality of single y-drivers(YDRVS) 110S is used for controlling the bitlines during write, read,and erase operation. Block YDRVS 110S will be described in detail belowin the description of the multilevel algorithm. Multiples of y-driverblock (YDRV) 110 are used for parallel multilevel page writing andreading to speed up the data rate during write to and read from themultilevel memory IC system 2000. A reference y-driver block (REFYDRV)116 including a plurality of single reference y-drivers (REFYDRVS) 116Sis used for the reference array block (MFLASHREF) 106. A redundanty-driver block (RYDRV) 112 including a plurality of single redundanty-drivers (RYDRVS) 112S is used for the redundant array (MFLASHRED) 102.The function of block (RYDRVS) 112S is similar to that of block (YDRVS)110S. A spare y-driver block (SYDRV) 114 including a plurality of singlespare y-drivers (SYDRVS) 114S is used for the spare array (MFLASHSPARE)104. The function of block (SYDRVS) 114S is similar to that of block(YDRVS) 110S. A page select block (PSEL) 120 is used to select onebitline out of multiple bitlines for each single y-driver (YDRVS) 110Sinside the block (YDRV) 110. Corresponding select circuit blocks forreference array, redundant array, and spare array are a reference pageselect block (PRSEL) 126, a redundant page select block 122, and a sparepage select block 124. A byte select block (BYTESEL) 140 is used toenable one byte data in or one byte data out of the blocks (YDRV) 110 ata time. Corresponding blocks for reference array, redundant array, andspare array are a reference byte select block 146, a redundant byteselect block 142, and a spare byte select block 144. The control signalsfor circuit blocks 116, 126, 146, 112, 122, 142, 114, 124, and 144 arein general different from the control signals for circuit blocks 110,120, and 140 of the regular memory array of the circuit block 100. Thecontrol signals are not shown in the figures.

A multilevel memory precision decoder block (MLMDEC) 130 is used foraddress selection and to provide precise multilevel bias levels overtemperature, process corners, and power supply as required forconsistent multilevel memory operation for the regular memory array ofthe circuit block 100 and for the redundant array 102. A multilevelmemory precision decoder block (MLMSDEC) 134 is used for addressselection and to provide precise multilevel bias levels overtemperature, process corners, and power supply as required forconsistent multilevel memory operation for the spare array 104.

An address pre-decoding circuit block (XPREDEC) 154 is used to providedecoding of addresses A<16:AN>. The term AN denotes the most significantbit of addresses depending on the size of the memory array. The outputsof block (XPREDEC) 154 couple to blocks (MLMDEC) 130 and block (MLMSDEC)134. An address pre-decoding block (XCGCLPRED) 156 is used to providedecoding of addresses A<11:15>. The outputs of block 156 also couple toblocks (MLMDEC) 130 and block (MLMSDEC) 134.

A page address decoding block (PGDEC) 150 is used to provide decoding ofaddresses A<9:10>. The outputs of block (PGDEC) 150 couple to blocks(PSEL) 120. A byte address decoding block (BYTEDEC) 152 is used toprovide decoding of addresses A<0:8>. The outputs of block (BYTEDEC) 152couple to blocks (BYTESEL) 140. An address counter block (ADDRCTR) 162provides addresses A<11:AN>, A<9:10>, and A<0:8> for row, page, and byteaddresses, respectively. The outputs of the block (ADDRCTR) 162 coupleto blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150, and (BYTEDEC)152. The inputs of the block (ADDRCTR) 162 are coupled from the outputsof an input interface logic block (INPUTLOGIC) 166.

The input interface logic block (INPUTLOGIC) 160 is used to provideexternal interface to systems off-chip such as the microcontroller 2001.Typical external interface for memory operation are read, write, erase,status read, identification (ID) read, ready busy status, reset, andother general purpose tasks. Serial interface can be used for the inputinterface to reduce pin counts for high-density chip due to a largenumber of addresses. Control signals 196L are used to couple theINPUTLOGIC 160 to the system microcontroller 2001. The INPUTLOGIC 160includes a status register that is indicative of the status of thememory chip operation such as pass or fail in program or erase, ready orbusy, write protected or unprotected, cell margin good or bad, restoreor no restore, etc. The margin and restore concepts are described morein detail in the multilevel algorithm description.

An algorithm controller block (ALGOCNTRL) 164 is used to handshake theinput commands from the block (INPUTLOGIC) 160 and to execute themultilevel erase, programming and sensing algorithms as needed formultilevel nonvolatile operation. The ALGOCNTRL 164 is also used toalgorithmically control the precise bias and timing conditions asrequired for multilevel precision programming.

A test logic block (TESTLOGIC) 180 is used to test various electricalfeatures of the digital circuits, analog circuits, memory circuits, highvoltage circuits, and memory array. The inputs of the block (TESTLOGIC)180 are coupled from the outputs of the INPUTLOGIC 160. The block(TESTLOGIC) 180 also provides timing speed-up in production testing suchas faster write/read and mass modes. The TESTLOGIC 180 is also used toprovide screening tests associated with memory technology such asvarious disturb and reliability tests. The TESTLOGIC 180 also allows anoff-chip memory tester to directly take over the control of variouson-chip logic and circuit bias blocks to provide various externalvoltages and currents and external timing. This feature permits, forexample, screening with external voltage and external timing or permitsaccelerated production testing with fast external timing.

A fuse circuit block (FUSECKT) 182 is a set of nonvolatile memory cellsconfigured at the external system level, at the tester, at the user, oron chip on-the-fly to achieve various settings. These settings caninclude precision bias levels, precision on-chip oscillator,programmable logic features such as write-lockout feature for portionsof an array, redundancy fuses, multilevel erase, program and readalgorithm parameters, or chip performance parameters such as write orread speed and accuracy.

A reference control circuit block (REFCNTRL) 184 is used to provideprecision reference levels for precision voltage levels as required formultilevel programming and sensing.

A redundancy controller block (REDCNTRL) 186 is for redundancy controllogic.

A voltage algorithm controller block (VALGGEN) 176 provides variousspecifically shaped voltage signals of amplitude and duration asrequired for multilevel nonvolatile operation and to provide precisevoltage levels with tight tolerance, as required for precisionmultilevel programming, erasing, and sensing.

A circuit block (BGAP) 170 is a bandgap voltage generator based on thebandgap circuit principle to provide a precise voltage level overprocess, temperature, and supply as required for multilevel programmingand sensing.

A voltage and current bias generator block (V&IREF) 172 is an on-chipprogrammable bias generator. The bias levels are programmable by thesettings of the control signals from the FUSECKT 182 and also by variousmetal options. A precision oscillator block (PRECISIONOSC) 174 providesaccurate timing as required for multilevel programming and sensing.

Input buffer blocks 196 are typical input buffer circuits, for example,TTL input buffers or CMOS input buffers. Input/output (io) buffer blocks194 includes typical input buffers and typical output buffers. A typicaloutput buffer is, for example, an output buffer with slew rate control,or an output buffer with level feedback control. A circuit block 196R isan open drained output buffer and is used for ready busy handshakesignal (R/BB) 196RB.

A voltage multiplier (also known as charge pump) block (VMULCKT) 190provides voltage levels above the external power supply required forerase, program, read, and production tests. A voltage multiplyingregulator block (VMULREG) 192 provides regulation for the block(VMULCKT) 190 for power efficiency and for transistor reliability suchas to avoid various breakdown mechanisms.

A flash power management block (FPMU) 198 is used to efficiently managepower on-chip such as powering up only the circuit blocks in use. TheFPMU 198 also provides isolation between sensitive circuit blocks fromthe less sensitive circuit blocks by using different regulators fordigital power (VDDD) 1032/(VSSD) 1033, analog power (VDDA) 1030/(VSSA)1031, and IO buffer power (VDDIO) 1034/(VSSIO) 1035. The FPMU 198 alsoprovides better process reliability by stepping down power supply VDD tolower levels required by transistor oxide thickness. The FPMU 198 allowsthe regulation to be optimized for each circuit type. For example, anopen loop regulation could be used for digital power since highlyaccurate regulation is not required; and a closed loop regulation couldbe used for analog power since analog precision is normally required.The flash power management also enables creation of a “green” memorysystem since power is efficiently managed.

Block diagram of the FPMU 198 is shown in FIG. 2B. A VDD 1111 and a VSS1000 are externally applied power supply and ground lines, respectively.A block (ANALOG POWER REGULATOR) 198A is an analog power supplyregulator, which uses closed loop regulation. The closed loop regulationis provided by negative feedback action of an operational amplifier (opamp) 1003 configured in a voltage buffer mode with a reference voltage(VREF1) 1002 on the positive input of the op amp 1003. A filtercapacitor (CFILL) 1004 is used for smoothing transient response of theanalog power (VDDA) 1030. A ground line (VSSA) 1031 is for analog powersupply. A block (DIGITAL POWER REGULATOR) 198B is a digital power supplyregulator, which uses open loop regulation. The open loop regulation isprovided by source follower action of a transistor 1006 with a referencevoltage (VREF2) 1005 on its gate. A pair of filter capacitor (CFIL4)1009 and (CFIL2) 1007 are used for smoothing transient response ofdigital power (VDDD) 1032. A loading element (LOAD1) 1008 is for thetransistor 1006. A ground line (VSSD) 1033 is for digital power supply.A block (IO POWER REGULATOR) 198C is an IO power supply regulator, whichuses open loop regulation similar to that of the digital power supply198B. The open loop regulation is provided by a transistor 1011 with areference voltage (VREF3) 1010 on its gate. A loading element (LOAD2)1013 is for transistor 1011. A pair of capacitors (CFIL5) 1014 and(CFIL3) 1012 are used for smoothing transient response of IO power(VDDIO) 1034. A ground line (VSSIO) 1035 is for IO power supply. A block198D includes various circuits that require unregulated power supplysuch as transmission switches, high voltage circuits, ESD structures,and the like.

A block (PORK) 1040 is a power on reset circuit which provides a logicsignal (PON) 1041 indicating that the power supply being applied to thechip is higher than a certain voltage. The signal (PON) 1041 istypically used to initialize logic circuits before chip operationbegins.

A block (VDDDET) 1050 is a power supply detection circuit, whichprovides a logic signal (VDDON) 1051 indicating that the operating powersupply is higher than a certain voltage. The block (VDDDET) 1050 isnormally used to detect whether the power supply is stable to allow thechip to take certain actions such as stopping the programming if thepower supply is too low.

A block (FPMUCNTRL) 1060 is a power supply logic controller, thatreceives control signals from blocks (PORK) 104, (VDDDET) 1050,(INPUTLOGIC) 160, (ALGOCNTRL) 164, and other logic control blocks topower up and power down appropriately power supplies and circuit blocks.The FPMUCNTRL 1060 is also used to reduce the power drive ability ofappropriate circuit blocks to save power. A line (PDDEEP) 1021 is usedto power down all regulators. Lines (PDAPOW) 1020, (PDDPOW) 1022, and(PDIOPOW) 1023 are used to power down blocks 198A, 198B, and 198C,respectively. Lines (PDDEEP) 1021, (PDAPOW) 1020, (PDDPOW) 1022, and(PDIOPOW) 1023 come from block (FPMUCNTRL) 1060.

It is possible that either closed or open loop regulation could be usedfor any type of power supply regulation. It is also possible that anypower supply could couple directly to the applied power supply (VDD)1111 without any regulation with appropriate consideration. For example,VDDA 1030 or VDDIO 1034 could couple directly to VDD 1111 if highvoltage transistors with thick enough oxide are used for analog circuitsor IO buffer circuits, respectively.

A typical memory system operation is as follows: a host such as themicrocontroller 2001 sends an instruction, also referred to as acommand, such as a program instruction via the CONTROL SIGNALS 196L andthe IO BUS 194L to the multilevel memory chip 2000 (see FIG. 1C). TheINPUTLOGIC 160 interprets the incoming command as a valid command andinitiates the program operation internally. The ALGOCNTRL 164 receivesthe instruction from the INPUTLOGIC 160 to initiate the multilevelprogramming algorithmic action by outputting various control signals forthe chip. A handshake signal such as the ready busy signal R/BB 196RBthen signals to the microcontroller 2001 that the multilevel memory chip2000 is internally operating. The microcontroller 2001 is now free to doother tasks until the handshake signal R/BB 196RB signals again that themultilevel memory chip 2000 is ready to receive the next command. Atimeout could also be specified to allow the microcontroller 2001 tosend the commands in appropriate times.

Read Operation:

A read command including a read operational code and addresses is sentby the microcontroller 2001 via the CONTROL SIGNALS 196L and IO BUS194L. The INPUTLOGIC 160 decodes and validates the read command. If itis valid, then incoming addresses are latched in the ADDRCTR 162. Theready busy signal (R/BB) 196RB now goes low to indicate that themultilevel memory device 2000 has begun read operation internally. Theoutputs of ADDRCTR 162 couple to blocks (XPREDEC) 154, (XCGCLPRED) 156,(PGDEC) 150, (BYTEDEC) 152, and (REDCNTRL) 186. The outputs of blocks154, 156, 150, 152, and 186 couple to blocks (MLMDEC) 130, (MLSMDEC)134, and block 100 to enable appropriate memory cells. Then theALGOCNTRL 164 executes a read algorithm. The read algorithm will bedescribed in detail later in the multilevel algorithm description. Theread algorithm enables blocks (BGAP) 170, (V&IREF) 172, (PRECISIONOSC)174, (VALGGEN) 176, and (REFCNTRL) 184 to output various precisionshaped voltage and current bias levels and algorithmic read timing forread operation, which will be described in detail later in thedescription of the multilevel array architecture. The precision biaslevels are coupled to the memory cells through blocks (MLMDEC) 130,(MLMSDEC) 134, and block 100.

In an embodiment, the read algorithm operates upon one selected page ofmemory cells at a time to speed up the read data rate. A page includes aplurality of memory cells, e.g., 1024 cells. The number of memory cellswithin a page can be made programmable by fuses, e.g., 512 or 1024 tooptimize power consumption and data rate. Blocks (PGDEC) 150, (MLMDEC)130, (MLMSDEC) 134, 100, and (PSEL) 120 select a page. All memory cellsin the selected page are put in read operating bias condition throughblocks (MLMDEC) 130, (MLMSDEC) 134, 100, (PSEL) 120, and (XCGCLPRED)156. After the readout voltage levels are stable, a read transfer cycleis initiated by the block (ALGOCNTRL) 164. All the readout voltages fromthe memory cells in the selected page are then available at they-drivers (YDRVS) 110S, (RYDRVS) 112S, and (SYDRVS) 114S inside block(YDRV) 110, (RYDRV) 112, and (SYDRV) 114, respectively.

Next, in the read transfer cycle the ALGOCNTR 164 executes a multilevelread algorithm to extract the binary data out of the multilevel cellsand latches them inside the YDRVS 110S, RYDRVS 112S, and SYDRVS 114S.This finishes the read transfer cycle. A restore flag is now set orreset in the status register inside the INPUTLOGIC 160. The restore flagindicates whether the voltage levels of the multilevel memory cellsbeing read have been changed and whether they need to be restored to theoriginal voltage levels. The restore concept will be described more indetail in the multilevel algorithm description. Now the ready busysignal (R/BB) 196RB goes high to indicate that the internal readoperation is completed and the multilevel memory device 2000 is ready totransfer out the data or chip status. The microcontroller 2001 now canexecute a status read command to monitor the restore flag or execute adata out sequence. The data out sequence begins with an external readdata clock provided by the microcontroller 2001 via the CONTROL SIGNAL196L coupled to an input buffer 196 to transfer the data out. Theexternal read data clock couples to the blocks (BYTEDEC) 152 and(BYTESEL) 140, 142, and 144 to enable the outputs of the latches insideblocks (YDRV) 110 or (RYDRV) 112 or (SYDRV) 114 to output one byte ofdata at a time into the bus IO<0:7> 1001. The external read data clockkeeps clocking until all the desired bytes of the selected page areoutputted. The data on bus IO<0:7> 1001 is coupled to themicrocontroller 2001 via IO BUS 194L through IO buffers 194.

Program Operation:

A program command including a program operational code, addresses, anddata is sent by the microcontroller 2001 via CONTROL SIGNALS 196L and IOBUS 194L. The INPUTLOGIC 160 decodes and validates the command. If it isvalid, then incoming addresses are latched in the ADDRCTR 162. The datais latched in the latches inside YDRV 110, RYDRV 112, and SYDRV 114 viablocks (BYTEDEC) 152, (BYTESEL) 140, 142, and 144, respectively. Theready busy signal (R/BB) 196RB now goes low to indicate that the memorydevice has begun program operation internally. The outputs of ADDRCTR162 couple to blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150,(BYTEDEC) 152, and (REDCNTRL) 186. The outputs of blocks 154, 156, 150,152, and 186 couple to blocks (MLMDEC) 130, (MLSMDEC) 134, and 100 toenable appropriate memory cells. Then the (ALGOCNTRL) 164 executes aprogram algorithm, which will be described in detail later in themultilevel algorithm description. The (ALGOCNTRL) 164 enables blocks(BGAP) 170, (V&IREF) 172, (PRECISIONOSC) 174, (VALGGEN) 176, and(REFCNTRL) 184 to output various precision shaped voltage and currentbias levels and algorithmic program timing for the program operation,which will be described in detail later in the description of themultilevel array architecture. The precision bias levels are coupled tothe memory cells through blocks (MLMDEC) 130, (MLMSDEC) 134, and block100.

In an embodiment, the program algorithm operates upon one selected pageof memory cells at a time to speed up the program data rate. Blocks(PGDEC) 150, (MLMDEC) 130, (MLMSDEC) 134, 100, and (PSEL) 120 select apage. All memory cells in the selected page are put in appropriateprogram operating bias condition through blocks (MLMDEC) 130, (MLMSDEC)134, 100, (PSEL) 120, and (XCGCLPRED) 156. Once the program algorithmfinishes, program flags are set in the status register inside the block(INPUTLOGIC) 160 to indicate whether the program has been successful.That is, all the cells in the selected page have been programmedcorrectly without failure and with enough voltage margins. The programflags are described more in detail in the multilevel algorithmdescription. Now the ready busy signal (R/BB) 196RB goes high toindicate that the internal program operation is completed and the memorydevice is ready to receive the next command.

Erase Operation:

An erase command including an erase operational code and addresses issent by the microcontroller 2001 via CONTROL SIGNALS 196L and IO BUS194L. The INPUTLOGIC 160 decodes and validates the command. If it isvalid, then incoming addresses are latched in the ADDRCTR 162. The readybusy signal (R/BB) 196RB now goes low to indicate that the memory devicehas begun erase operation internally. The outputs of ADDRCTR 162 coupleto blocks (XPREDEC) 154, (XCGCLPRED) 156, (PGDEC) 150, (BYTEDEC) 152,and (REDCNTRL) 186. The outputs of blocks 154, 156, 150, 152, and 186couple to blocks (MLMDEC) 130, (MLSMDEC) 134, and 100 to enableappropriate memory cells. Then the ALGOCNTRL 164 executes an erasealgorithm. The ALGOCNTRL 164 enables blocks (BGAP) 170, (V&IREF) 172,(PRECISIONOSC) 174, (VALGGEN) 176, and (REFCNTRL) 184 to output variousprecision shaped voltage and current bias levels and algorithmic erasetiming for erase operation. The shaped voltage for erase is to minimizeelectric field coupled to memory cells, which minimizes the damage tomemory cells during erasing. The precision bias levels are coupled tothe memory cells through blocks (MLMDEC) 130, (MLMSDEC) 134, and block100.

In an embodiment, the erase algorithm operates upon one selected eraseblock of memory cells at a time to speed up the erase time. An eraseblock includes a plurality of pages of memory cells, e.g., 32 pages. Thenumber of pages within an erase block can be made programmable by fusesto suit different user requirements and applications. Blocks (PGDEC)150, (MLMDEC) 130, (MLMSDEC) 134, 100, and (PSEL) 120 select a block.All memory cells in the selected block are put in erase operating biascondition through blocks (MLMDEC) 130, (MLMSDEC) 134, 100, (PSEL) 120,and (XCGCLPRED) 156. Once the erase algorithm finishes, the erase flagsare set in the status register inside the block (INPUTLOGIC) 160 toindicate whether the erase has been successful. That is, all the cellsin the selected page have been erased correctly to desired voltagelevels without failure and with enough voltage margins. Now the readybusy signal (R/BB) 196RB goes high to indicate that the internal eraseoperation is completed and the multilevel memory device 2000 is ready toreceive the next command.

Multilevel Array Architecture:

The demanding requirements associated with putting together a billiontransistors on a single chip with the ability to store multipleprecision levels per cell and operating at a very high speed arecontradictory. These requirements need innovative approaches and carefultradeoffs to achieve the objective. Examples of tradeoffs and problemswith prior art implementation are discussed below. In conventional priorart architectures, a voltage drop along a metal line of a few tens ofmillivolts could be easily tolerated. Here, in a super high densitynonvolatile multilevel memory integrated circuit system such a voltagedrop can cause unacceptable performance degradation in precision levelsdue to the high number of levels stored per memory cell. In conventionalarray architectures, a bit line capacitance in the order of 10 picofarads would be a non-issue. Here it may be unworkable due to the highdata rate required. In prior art array architectures a bias levelvariation from one memory cell to another in the order of +/−30 percentwould be a typical situation. Here such a bias variation would be aserious performance problem. In prior art array architectures, the totalresistance of a memory source line in the order of a few hundreds ofohms would be a typical situation, here a few tens of ohms is a seriousproblem. The huge number of memory cells of the giga to tera bithigh-density memory system compounds the matter even further by makingthe memory source line longer. Another challenge facing the multilevelsystem is maintaining high speed sensing and programming with low power,again requiring tradeoffs. Another challenge facing the multilevelsystem is high speed sensing and programming with very high precisionvoltages due to a high number of levels stored per digital multilevelmemory cell, again a conflicting demand. Another challenge facing themultilevel system is high speed sensing and programming consistentlyevery time over many years, process corners, temperature, and powersupply variation.

To get an appreciation of the order of magnitude of the difficultyinvolved in the super high density multilevel nonvolatile memory system,numerical examples will be given corresponding to a one giga bit arrayarchitecture system suitable for 256 levels, i.e., 8 bits. The array isthen organized as 8192 bitlines or columns and 16384 rows or wordlinesfor a total of 134,217,730 physical cells.

One sensing level, V1level, =multilevel sensing range/2^(N), N=number ofdigital bits stored per memory cell. Multilevel sensing range is thereadout voltage range from sensing a multilevel memory cell. Assumingthe multilevel sensing range from the multilevel memory cell availableis 2048 millivolts, then V1level=2048/256=8 millivolts.

A very high data rate is required for applications such as image or highdensity data storage. For example, write and read rates of a mega byteper second are required. To achieve this high data rate, parallelwriting and sensing is required for the super high density nonvolatilemultilevel memory integrated circuit system. In the present embodiment,a total of 1024 y-drivers (YDRVS) 110S inside blocks (YDRV) 110 areused. This allows 1024 memory cells to be written and sensed at the sametime in a page mode manner, effectively increasing the speed by a factorof 1024 over single cell operation. The number of bitlines multiplexedinto one single y-driver (YDRVS) 110S is =8192/1024=8 bitlines.

A program algorithm described in more detail elsewhere in thisspecification is able to achieve desired multilevel resolution. The reador program multilevel resolution is the smallest voltage range in reador program, respectively, needed to operate the multilevel memory cellscorrectly. An erase algorithm first erases the memory cells to make thecell readout voltage reaching a certain desired voltage level. Then theiterative program algorithm is applied to the memory cells. The programalgorithm includes a plurality of verify-program cycles. Averify-program cycle includes a verify cycle followed by a programcycle. A verify cycle is done first to inhibit the cell from the firstprogramming pulse if the cell is verified, therefore preventing possibleover-programming. Over-programming means that after a programming pulsethe cell sensing level passes a desired voltage level by more than adesired voltage amount. A verify cycle is used to determine whether thedesired readout sensing level has been reached. If the desired readoutsensing level is reached, the cell is inhibited from furtherprogramming. Otherwise, the cell is enabled for the next program cycle.A program cycle is used to change incrementally the charge stored in thecell and the corresponding cell sensing readout voltage. Instead of averify-program cycle, a program-verify cycle can be used. Aprogram-verify cycle begins with a program cycle followed by a verifycycle. In this case, care should be taken to ensure that the firstprogramming pulse does not cause over-programming.

In an embodiment the program cycle includes applying a voltage on thesource line, (interchangeably referred to as common line [CL]) (VCL),with a predetermined program pulsewidth (TPPWD) and a predeterminedprogram bias cell current (Ipcell). The verify cycle makes use of thevoltage mode sensing as shown in FIG. 2C, which applies a referencevoltage (VCLRD) on the source line (CL), another reference voltage(VCGRD) on the control gate, and a predetermined read bias current(Ircell) on the bitline and through the memory cell. The current(Ircell) is applied to the bitline and the memory cell through selecttransistors which are not shown. The resulting voltage on the bitline isthe sensing readout voltage (VR), which has a unique relationship to thecharge on the floating gate. The voltage mode sensing is also usedduring read. To change incrementally the readout sensing voltage to thenext value (VR+dVR), with dVR equals to the incremental readout sensingvoltage change, the next program cycle is repeated with the common linevoltage increased incrementally to (VCL+dVCLP), with dVCLP equals to theincremental programming voltage change.

The number of verify-program cycles (NC) is dependent on the number ofvoltage levels and various margins of the memory system. For example,for an equivalent 8-bit digital multilevel cell, there are 2^(N)=2⁸=256levels, with N=8. The minimum possible number of verify-program cycles(NC) required would be 256. To cover variations due to cell-to-cellvariation, temperature, process corners, an algorithm may require, forexample, approximately 1.4×256=360 verify-program cycles. To covervarious margins needed such as for data retention and programmingdistribution, the number of verify-program cycles required is actuallyhigher. Assuming a factor of 2 due to various margin coverage, thenumber of verify-program cycles is approximately equal to 720. The exactnumber of verify-program cycles is typically varied depending on variousmemory technologies and particular desired performance targets.

For write data rate of 1 mega byte per second and for 8-bit digitalmultilevel operation with 1024 bytes per page, the write timing per pageis, TWRT=# of bytes written in parallel/data rate=1024 bytes per page/1mega bytes/second=1024 μs=1.024 ms per page.

Hence the time to execute each program-verify cycle (TPV) must be lessthan TWRT/NC=1.024 ms/720=1.42 μs. This fast timing coupled withparallel operation of 1024 cells has important implication on memorycell program speed, capacitance loading, power consumption and othereffects as will be described below.

Typical process parameters of a sub-micron memory cell are as follows. Atypical diffused source line resistance per cell is 100 ohms. A typicalbitline resistance per cell is 80 milliohms. A typical silicided rowline resistance per cell is 20 ohms. A typical source line capacitanceper cell is 2 fF. A typical bitline capacitance per cell is 1.5 fF. Anda typical row line capacitance per cell is 3 fF.

Hence for the 8192×16384 array, the total bitline capacitance isCBL=˜6384×1.5 fF=25 pF, where “=˜” is defined as approximately equal to.The total metal bitline resistance RBL=˜16384×0.08=1330 ohms. The totaldiffused source line resistance is RSL=8192×100=819 K ohms. The totalrow line resistance is RWL=8192×20=164 K ohms. For a typical memorysystem, the diffused source line is strapped by metal along the sourceline, with approximately 80 milliohms per cell, in this caseRSL=8192×0.08=655 ohms.

In conventional stacked gate drain-side CHE programming (abbreviated asCHE flash program), the single cell current is typically 1 ma, whichcauses a voltage drop along a single metal bitline of =˜1 ma×RBL=˜1ma×1330 ohms=1330 millivolts, which is unacceptable since it is muchgreater than 1 level=8 millivolts. In SSI flash programming (abbreviatedas SSI flash program), the typical cell current can be lowered to 1 μa,which causes a voltage drop along a single metal bitline of =˜1 μa×1330ohms=1.33 millivolts, which is acceptable.

For 1024 cells drawing the cell current (Icell) continuously, thevoltage drop (DVCL) along the source line from the driver to the otherend follows the geometric equation:DVCL=0.5*P*(P+1)*R8cell*Icell,tm  (1)where R8cell=the metal source line resistance for 8 cells in series=0.08ohms×8=0.64 ohms, and P=1024.

Along the source line, for 1024 cells programming simultaneously, thetotal current is 1024×1 ma=1.024 A for the CHE flash program and =1024×1μa=1.024 ma for the SSI flash program. The power needed for the drainside CHE flash programming for parallel page mode operation isunsustainable due to very high current. Additionally, the voltage dropalong the metal source line by equation (1) is =˜0.5×1024*1025*0.64*1ma=336 Volts for CHE. This is unworkable for CHE flash technology.Similarly, the source line voltage drop for the SSI flash =˜336millivolts. This is also unworkable in the multilevel program for thefollowing reasons.

For a multilevel nonvolatile system, in one program cycle, the cellsensing voltage can only shift (dVR) a maximum of <(Q*V1level) forreliable sensing, where Q was 0.5 in the prior example. However Q couldvary from ⅓ to ⅛ for long term reliability. This is needed, for example,to allow for sensing margin, verify margin, program disturb, dataretention, and endurance. The number of cells programming simultaneouslywithin a selected page can vary between as many as 1024 to as few asonly one from one program cycle to the next. Thus the total programcurrent flowing through the common line CL could change by a factor of1024 from one program cycle to the next. The resulting worst casevoltage change in the source line VCL from one program cycle to the nextis dVCL=˜336 millivolts for SSI flash. This voltage jump in VCL causesthe only remaining programming cell to over program, which causes thecell sensing voltage to shift much greater than the (Q*V1level). Hence,the challenge is to bring the voltage drop dVCL to an acceptable levelduring programming.

For verifying after programming multilevel memory cells, conventionalmethods would shut off the read cell currents for cells that havealready reached their desired verifying levels, this would cause thevoltage shift dVCL in verify as much as in programming as describedabove. This voltage jump dVCL would couple to the memory cells and causea large jump in cell sensing voltage. This undesired large jump in cellsensing voltage causes an error in sensing, herein called a sense errorVRerr. This sense error should be much less than (Q*V1level). Hence thislarge jump is unacceptable. The invention solves the problem by enablingthe total current all the time whether the cells have been verified ornot. This mitigates the change in the source line voltage. However a newproblem surfaces as compared to that in programming. As temperaturechanges from −45 C to +85 C the resistance of the source line metal linechanges by about 40%, hence the source line voltage drop changes byabout 40%, which causes an additional sense error VRerr in read. Thissense error should be much less than (Q*V1level) to prevent overall readmargin degradation. Therefore, an array architecture is needed toachieve this, as will be described in detail below.

With 1024 cells operating simultaneously, assuming sense currentIrcell=10 μa, the total sense current is =˜1024×10 μa=10.24 ma flowinginto the source line. This presents several problems. With powerspecification for a typical memory chip ICC=20-30 ma. This 10.24 ma is abig percentage of the power specification. To deliver 10.24 ma whilemaintaining a precise voltage level VCLRD, VCLRD is defined as thevoltage in read on CL line, requires a challenging decoding and driverscheme, which will be addressed in the description of the multileveldecoding scheme. Large current flowing across the source line alsocauses the voltage drop as described above.

High data rate, meaning high sense speed and write speed, is requiredfor data intensive application. The speed is proportional to capacitanceand voltage swing and inversely proportional to the current,T=C*V/I  (2).

For typical bitline capacitance as calculated above, CBL=25 pF andassuming voltage swing V=1V, and assuming available current I=10 μa, thetime it takes to charge or discharge a bitline as needed in verify orprogram cycle is, TBL=25 pF*1V/10 μa=2.5 μs. This is greater than theTPV=1.42 μs as calculated above. At least a 2× or better timing isrequired for TBL to allow for various settling time, sensing time, andprogramming time. Increasing the current would cause higher powerconsumption, large decoding driver, and voltage problems as describedabove.

Further, in programming 1024 cells in parallel, the programming currentis supplied from an on-chip voltage multiplier, also known as a chargepump. The on-chip voltage multiplier multiplies the low voltage powersupply, e.g., 2.5 V to the required higher voltages. Allowing areasonable area penalty from the on-chip voltage multiplier, a totalcurrent of 100 μa is allowed for programming. The programming currentper cell is 100 μa/1024=0.1 μa. This causes a TBL=25 pF*1V/0.1 μa=250μs, which is even more severe of a timing problem. Here an improvementof more than 2 order of magnitude or better in speed is needed. Theinvention describes array architectures with suitable operating methodsto achieve this improvement and will be described below.

FIG. 3A is the block diagram of a super high-density digital nonvolatilemultilevel memory array architecture which is capable of >8-bitmultilevel operation. The block 100 has been expanded from FIG. 2A toshow the sub-blocks inside. A multilevel precision memory decoderMLMDECS 132 is used for delivering bias voltage levels with tighttolerance over temperature, process, and power supply variation formultilevel memory cells. A multilevel memory sub-array MFLSUBARY 101includes a plurality of single multilevel memory cells. Other blocks inFIG. 3A have already been described in association with the descriptionof FIG. 2A.

A block (PSEL) 120 includes a plurality of circuit blocks (PSELS) 120S.FIG. 3B shows details of a page select circuit (PSELS) 120S that selectsa pair of bitlines at a time. Transistors 120A-D are select transistors.Transistors 120E-H are inhibit transistors. Lines (P0) 120K, (PP1) 120M,(PP2) 120O, and (PP3) 120Q are complementary signals of lines (PP0B)120L, (PP1B) 120N, (PP2B) 120P, and (PP3B) 120R, respectively. Line(BLYDRV) 120Y goes to one y-driver (YDRVS) 110S inside the block (YDRV)110. Block (YDRVS) 110S will be described in detail later in thedescription of the multilevel algorithm. Lines (BLTP0) 240P, (BLTP1)241P, (BLTP2) 242P, and (BLTP3) 243P couple to the bitlines in block 101and couple to a set of lines (BLP0) 240, (BLP1) 241, (BLP2) 242, and(BLP3) 243 of the circuit block 290 in FIG. 4A.

FIG. 3C shows a block diagram of a block (MFLSUBARY) 101. A block(MFLSUBARY) 101 includes a plurality of blocks (ARYSEG0) 290. Blocks(ARYSEG0) 290 are first tiled horizontally NH times and then thehorizontally tiled blocks 290 are tiled vertically NV times. For a pagewith 1024 memory cells, NH is equal to 1024. NV is determined such thatthe total number of memory cells is equal to the size of the desiredphysical memory array.

FIG. 4A shows a basic array unit (ARYSEG0) 290. A block (RD1SEG) 300 isa multilevel decoding block. A plurality of the blocks RDLSEG makes upthe circuit block (MLMDEC) 130. In the block (ARYSEG0) 290, there are 8columns and FIG. 4A shows only 8 rows of memory cells, while other rows,e.g., 120 rows, are not shown for clarity. Each ARYSEG0 290 includes aplurality, e.g. 8, of array blocks (ARYLBLK) 290A tiled vertically. Aset of transistors 220, 221, 222, 223, 224, 225, 226, 227 couplesrespectively a set of segment bitlines (SBLO) 240A and (SBL1) 240B,(SBL2) 241A and (SBL3) 241B, (SBL4) 242A and (SBL5) 242B, (SBL6) 243Aand (SBL7) 243B to a set of top bitlines (BLP0) 240, (BLP1) 242, (BLP2)242, and (BLP3) 243, respectively. Top bitlines refer to bitlinesrunning on top of the whole array and running the length of theMFLSUBARY 101. Segment bitlines refer to bitlines running locally withina basic array unit ARYSEG0 290. A set of transistors 230, 231, 232, 233,234, 235, 236, 237 couples respectively segment bitlines (SBL0) 240A and(SBL1) 240B, (SBL2) 241A and (SBL3) 241B, (SBL4) 242A and (SBL5) 242B,(SBL6) 243A and (SBL7) 243B to an inhibit line (VINHSEGO) 274. A line(CL0) 264 is the common line coupled to common lines of the first fourrows of memory cells. A line (CL3) 269 couples to common lines of thelast four rows of memory cells. A set of control gates (CG0) 262, (CG1)263, (CG2) 265, (CG3) 266 couples to control gates of memory cells ofthe first four rows respectively. A set of control gates (CG12) 267,(CG13) 268, (CG14) 270, (CG15) 271 couples to control gates of memorycells of the last four rows, respectively. A pair of inhibit selectlines INHBLB0 272 and INHBLB1 273 couples to gates of transistors 231,233, 235, 237 and transistors 230, 232, 234, 236 respectively. A pair ofbitline select lines (ENBLB0) 260 and (ENBLA0) 261 couples to gates oftransistors 221, 223, 225, 227 and transistors 220, 222, 224, 226,respectively.

Multiple units of the basic array unit (ARYSEG0) 290 are tiled togetherto make up one sub-array (MFLSUBARY) 101 as shown in FIG. 3C. Andmultiples of such (MFLSUBARY) 101 are tiled horizontally to make up thefinal 8192 columns for a total of 32768×8192=268,435,460 physical memorycells, or called 256 mega cells. The logical array size is 256 megacells×4 bits per cell=1 giga bits if 4-bit digital multilevel memorycell is used or 256 mega cells×8 bits per cell=2 giga bits if 8-bitdigital multilevel memory cell is used. The top bitlines (BLP0) 240,(BLP1) 241, (BLP2) 242, and (BLP3) 243 run from the top of the array tothe bottom of the array. The segment bitlines (SBL0) 240A, (SBL1) 240B,(SBL2) 241A, (SBL3) 241B, (SBL4) 242A, (SBL5) 242B, (SBL6) 243A, and(SBL7) 243B only run as long as the number of rows within a segment, forexample, 128 rows. Hence the capacitance contributed from each segmentbitline is very small, e.g., 0.15 pF.

The layout arrangement of the top bitlines 240-243 in relative positionwith each other and with respect to the segment bitlines (SBL0) 240A,(SBL1) 240B, (SBL2) 241A, (SBL3) 241B, (SBL4) 242A, (SBL5) 242B, (SBL6)243A, (SBL7) 243B are especially advantageous in reducing the bitlinecapacitance. The purpose is to make the top bitlines as truly floatingas possible, hence the name of truly-floating-bitline scheme.

In an embodiment a shown in FIG. 5A, lines 240, 241, and 242 are in themiddle, sandwiched between lines 240A, 240B, 241A and 241B in the bottomand lines (CL0) 264 in the top. Furthermore, line 240 is on top of thespacing between lines 240A and 240B and line 241 is on top of thespacing between lines 241A and 241B. This has the benefit of reducingsignificantly the bottom plane capacitance of line 240 and line 241since the oxide below each line is almost doubled. The lines 240 and 241could be positioned on top of lines 240A and 241A, respectively, whenthe sidewall capacitance reduction outweighs the benefit of the bottomplane capacitance reduction. The sidewall capacitance refers to thecapacitance resulting from the vertical walls of a line, the bottomplane capacitance refers to the capacitance from the bottom of a line,and the top plane capacitance refers to the capacitance from the top ofa line.

In another embodiment, as shown in FIG. 5B, the top bitlines 240-242have been positioned all the way to the top metal of a multi-layer metalintegrated circuit system. For example, for a 5-layer metal integratedcircuit system, the top bitlines are metal 5 layer. This avoids the topplane capacitance of the top bitlines 240-242. This also reduces thebottom plane capacitance of the top bitlines 240-242 by a factor of asmuch as 4 if metal 5 is used. The reduction factor of 4 is due to theoxide below the line increasing by a factor of about as much as 4. Alsosince the top bitlines 240-242 are spaced further apart as compared tothe segment bitlines, the sidewall capacitance is reduced significantly.The top bitlines are now almost floating on top of the array. The endeffect is more than on order of magnitude reduction in bitlinecapacitance. Also since the top bitlines 240-242 spacing are relaxed,the width of the top metal lines can be made larger to reduce the metalbitline resistance.

The reduction in bitline capacitance results in a corresponding increasein speed. To help increase the speed in programming, abitline-stabilization-assisted operating method can be applied and isdescribed as follows. At the beginning of the programming cycle, abitline stabilization control signal is used to set all the bitlines toa predetermined voltage VBLPRE, e.g., 0.4-0.8 V. Then high voltage VCLis applied to selected memory common lines for programming. Now thebitlines only have to move partially to a final voltage. This speeds upthe TBL timing.

There is an important transient effect related to bitline capacitance inprogramming. For high speed writing, each program cycle takes time inthe microsecond range. The program bias condition for a memory cell iscontrol gate voltage VCGP, =˜0.7-2.5 V, bitline cell current Ipcell,=˜50-500 nA, and common line voltage VCL going from a low, =˜0 V, to ahigh programming voltage, =˜8-13 V. As the VCL ramps from a low to ahigh voltage, there is a transient current flowing through the memorycell to charge up the bitline node capacitance. This transient currentflowing through the cell contributes to the cell programming in additionto the programming current Ipcell. Prior art CHE programming would notbe bothered with this effect since the additional transient programmingcurrent is small compared to the actual programming current. However,for a very fine programming voltage level control as required for highbits per cell, this effect will cause the programming level to beuncontrollable, making the multilevel memory system useless. Thefollowing example is given to appreciate the magnitude of this transientcurrent. Assuming program VCL ramp time=1 μs, CBL=1 pF, the voltage thebitline has to slew=1 V, then, by equation (2), I=CV/T=1 pF×1 V/1 μs=1μA, which can be 10× the programming current. Hence a method is neededto reduce the transient programming current.

Two approaches are shown in FIG. 5C to reduce this transient phenomenon.In one embodiment, 2-step ramp rate control approach greatly reducesthis transient effect without prolonging the programming time asfollows. First VCL ramps fast during TRP1 to an intermediate voltageVCLINT, e.g., 2-6 V, then VCL stays at an intermediate voltage for afinite time TVCLINT, then VCL ramps slow during TRP2 to a final voltageVCLFIN. The first fast ramp with the flat intermediate time TVCLINT willlet transient current flowing through the cell to stabilize most of thecell capacitances such as CBL in a short time and at sufficiently lowVCL voltage so that insignificant programming takes place while thetransient current is flowing. The TRP1 is made fast to consume littleprogramming time. The second slow ramp then brings the cell to a finalprogramming voltage without affecting the programming rate since verylittle current is flowing through the cell in the second ramp.

Another embodiment of the ramp rate control is a fast-slow ramp ratecontrol approach. VCL first ramps fast during TRP1 to an intermediatevoltage VCLINT, then VCL ramps slow during TRP2 to a final voltageVCLFIN. The first ramp TRP1 is faster than that of the TRP2 ramp toallow the transient current during the first ramp TRP1 to stabilizequickly all the cell capacitances while VCL is low enough to not causesignificant programming.

The ramp rate can be generated by a RC network, meaning the rate iscontrolled by a certain capacitance multiplied by a certain resistance,or by a CV/I network, meaning the rate is controlled by a certaincapacitance multiplied by a voltage swing divided by a certain biascurrent. Further, the ramp rate can be programmable by programmablefuses as a function of bitline capacitance to optimize the programmingtime without introducing adverse transient current. That is the ramprate is made to be faster for smaller bitline capacitance.

The common line CL0 264 is common to four rows of memory cells for thefollowing reason. Allowing 4 mV voltage drop along the CL line duringprogramming to avoid programming error as described previously, with1024 cells operating simultaneously with 0.1 μa drawn per cell, thevoltage drop by equation (1) is, dVCLP=4 mV=0.5*(1024) (1025) R8cell*0.1μa, hence R8cell=76 milliohms. For a typical CL line with the line widthhalf as wide as the memory cell, the CL resistance per cell is =˜80milliohms, for 8 cells in series, R8cell is 8×80=640 milliohms, which ismuch greater than 76 milliohms. Hence by making CL line 264 four memorycells wide, R8cells is =˜80 milliohms. The reason the width of the lineCL 264 cannot be made arbitrarily large is due to the program disturb.As the high voltage is applied to CL line 264 in programming, all thecells connected to the CL line 264 will see the VCL voltage whether theyare selected for programming or not. The more cells connected to thesame CL line, the longer time for the disturb for the unselected cells.

Shown in FIG. 4A are the metal strapping lines (CL0STRAP) 264S and(CL3STRAP) 269S of the common lines that connect the diffusion commonlines to the metal common lines. The metal strapping could be done every8, 16, or 32 memory cells depending on an allowable voltage drop alongthe common line diffusion inside the strapping. This voltage dropdepends on the diffusion common line resistance for a given operatingcurrent.

An alternative method that mitigates the voltage drop problem along thecommon line in the program cycle is by theconstant-total-current-program scheme. Namely by keeping the same totalcurrent flowing all the time independent of whether the cells have beenverified or not, the common line voltage drop is kept constant duringprogramming. This could be done for example, by adding additionalswitching transistors in the array every 8, 16, 32, or 64 memory cellsand switching into the CL line the current equivalent to the currentfrom verified cells.

Table 1 shows the operating conditions for the memory array in read,erase, and program. The array operating conditions are shown for thecell 200 of the block ARY1BLK 290A in FIG. 4A, of a selected page forread and program. The selected cell 200 is one cell out of 1024 selectedcells within a selected page. The other 1023 selected cells belong tothe other 1023 ARYSEG0 290 connected horizontally. The array operatingconditions are also shown for all cells connected to CL0 264 for erase.

As shown in Table 1, the operating conditions are such that all theunselected memory cells see no voltage other than 0 volts. This reducessignificantly the power consumption. This is also particularlyadvantageous for improved speed in very high-density memory chips sinceall the necessary driver circuits only see the loading from the selectedmemory cells. The loading from the whole array is tremendous due tolarge number of transistors in array, e.g., 256 million transistors,with its tremendous diffusion, metal and poly interconnect parasitics.For example, one bitline capacitance, CBL is 25 pF, with 8192 bitlinesthe total bitline capacitance is 8192×25 pF=204 nF. This would require atremendous amount of power during signal switching, for example, toinhibit all the bitlines during programming. Also not shown in Table 1,the unselected control signals ENBLAs, ENBLBs, INHBLAs, and INHBLBs forunselected array units ARYSEG0 290 only see 0 or VDD but not themultiplied high voltage. This again saves significant power andincreases speed due to no loading from unselected control circuits.

Another factor that is reduced greatly is the excessive leakage currentfrom the bitline to ground due to junction leakage, bitline to bitlineleakage, band-to-band tunneling, and cell subthreshold conduction. Forexample, for a typical leakage of 10 pA per cell, with 16,384 cells perbitline, the total leakage is 164 nA, which is greater than Ipcell=100nA. This implies that the multilevel programming will be uncontrolleddue to the uncontrollable excessive leakage current contributing to thecontrolled programming current Ipcell. With the inhibit and segmentationscheme, the total leakage current is reduced to 128×10 pA=1.28 nA, whichis much less than Ipcell=100 nA.

FIG. 4B shows an alternative array architecture in which the decodedinhibit line VINHSEGO1 274B is shared between any two adjacent segments.This has the benefit of reducing the number of inhibit lines in thearray.

FIG. 4C shows an alternative array architecture in which the inhibitline VINH 999 is shared for all the segments. This has the benefit ofsharing one inhibit line for the whole array.

FIG. 4D shows an alternative array architecture in which a set ofinhibit select line INHBLA1-3 and INHBLB1-3 275 to 280 are used toinhibit all segment bitlines except the selected segment bitline. VINH999 is shared for all the segments. The operating method makes use of asegment cascading scheme that is described as follows. To even isolatethe bitline capacitance further, bitline select transistors 220-227 arealso used as cascading transistors in programming in addition to theselect and inhibit function. In programming, cell 200 for example, thevoltage on line 261 is initially pulsed high to pass inhibit voltageVINH 999 from a page select (PSELS) 120S into the selected segmentbitline (SBL0) 240A. Then the voltage on line ENBLA0 261 is pulsed to acascading voltage (VPBCAS), e.g., 1 V. A precharge signal then chargesthe selected top bitline (BLP0) 240 to 0.3V. The final voltage on thetop bitline (BLP0) 240 is =˜0.3 V since 1V−VT=˜0.3 V. Hence the voltageon line BLP0 240 no longer changes during programming. The voltage onthe segment bitline, however, still changes as VCL is applied andstabilized. But the capacitance on the segment bitline is minimal,=˜0.15 pF. Here the operating method just described could also apply tothe array shown in FIG. 4A but the inhibit voltages on the unselectedsegment bitlines are floating. The array shown in FIG. 4D just makessure all the unselected segment bitlines are kept at a constant inhibitvoltage (VINH) 999.

FIG. 4E shows another array suitable for the method just describedabove. It needs a set of 4 additional lines (INHBLAB0-3) 281-284 and aset of 8 additional transistors 240I-247I for inhibit decoding. Howeveradditional transistors 240I-247I occupy less die area than that requiredfor additional inhibit decoding lines 275-280 in FIG. 4D.

FIG. 4F shows an array architecture similar to that in FIG. 4A with theinhibit transistors physically at the top of the segment array.

Note that it is possible to do one top bitline per one segmented bitlinein the ARYSEG0 290. In this case, the sidewall capacitance from one topbitline to adjacent top bitlines increases due to reduced spacingbetween the top bitline and the adjacent top bitlines.

Note that it is also possible to do one top bitline per more than twosegmented bitlines in the ARYSEG0 290. In this case, more decodingtransistors are needed in the array to select one segmented bitline outof more than two segmented bitlines, which leads to more die size.However the sidewall capacitance from one top bitline to adjacent topbitlines decreases due to increased spacing between the top bitline andthe adjacent top bitlines. This reduction of capacitance may not besignificant if the spacing is already wide enough.

An alternative embodiment of reducing the bitline capacitance is byhierarchical interconnect segmentation that is an extension over theprevious concept as follows. A first segment bitline running in firstlayer of metal couples to a plurality of memory cells. A second segmentbitline running in second layer of metal is coupled to a plurality offirst segment bitlines by bitline segment transistors through viasbetween metal 1 and metal 2. Third segment bitline running in thirdlayer of metal is coupled to a plurality of second segment bitlines byother bitline segment transistors through vias between metal 1 and metal2 and metal 3. This can continue to higher metal layers. This approachallows optimization of horizontal spacing, vertical spacing,interconnect width, and interconnect length between different layers ofinterconnect metals for minimum capacitive coupling between metalinterconnect lines. This results in further reduced bitline capacitance.TABLE 1 Array Operating Conditions READ ERASE PROGRAM SELECTED SEGMENTS:CG0 3-6 V 8-13 V 0.7-2.5 V CG1, 2, 3 0 8-13 V CG4-15 0 0 0 Rest of all 00 0 CG lines CL0 2-3 V 0 4-13 V CL1, 2, 3 0 0 0 Rest of all 0 0 0 CLlines BL0, 8, 16 . . . 0 TO 2-3 V FL or 0 V 0-0.8 V BL1-7, 9-15, VINHVINH VINH 17-23, . . . UNSELECTED SEGMENTS: All CG lines 0 V 0 V 0 V AllCL lines 0 V 0 V 0 V All BL lines 0 V 0 V 0 VMultilevel Memory Decoding:

FIG. 6 shows the block diagram of the multilevel decoding scheme. Theinvention provides precision voltages with millivolt control tolerancesto the memory array over temperature, process corners, and power supplyvariation. The invention provides these voltages in an efficient manner,meaning deliver power where it is needed and reducing the output loadingthrough circuit configuration. The invention also provides a multilevelprecision decoding circuit with minimum area overhead.

As discussed in the array architecture section, the voltage drop alongthe common line would cause a programming error as well as sense errorin read. Hence the drop is brought down to a manageable level. Bypartitioning a common line into small line sections, with drivers onboth sides of each of the line sections, the voltage drop is reduced.However, prior art partition would cause a tremendous area penalty dueto the large amount of decoding lines and circuits. This inventionprovides an enhanced decoding circuit by routing the interconnect in thehigher metal layers and by using circuit configurations suitable formultilevel decoding.

The block (VCGCLPRED) 156 has been expanded to include sub-blocksinside. Common line predecoder and driver (XCLPREDRV) 950 providepredecoded common lines with precision voltages to regular memory commonlines in block 130 and 132. A common line predecoder and driver(XCLSPREDRV) 954 provides predecoded common lines with precisionvoltages to spare memory common lines in block 134. The circuit block954 is functional equivalent to circuit 950. A control gate predecoder(XCGPREDEC) 951 provides predecoded control gate lines to block 130. Aspare control gate predecoder (XCGSPREDEC) 952 provides predecodedcontrol gate lines to block 134. A bitline predecoder (BLXDEC) 953provides predecoded bitlines to block (MLMDEC) 130. All other circuitblocks have been described in association with FIG. 2A.

FIG. 7 shows one segmented decoder (RD1SEG) 300. The RD1SEG 300 selectsor deselects a plurality of basic array unit (ARYSEG0) 290 connectedhorizontally. The RD1SEG 300 includes a circuit segmented supply decoder(RDSGPSDEC) 301, a segmented bitline decoder (RDSGBLDEC) 302, asegmented common line pre-decoder (RDSGCLPDEC) 302B, a segmented inhibitdecoder (RDSGINHDEC) 303, and multiples of a sub-block decoder(RD1SUBBLK) 304. The RDSGPSDEC 301 decodes the high voltage supply foreach segmented decoder (RDLSEG 300). The high voltage supplies for theunselected segmented decoders (RD1SEG) 300 are disabled and hence poweris minimized due to much less loading and die size is reduced due to asmaller voltage multiplier. The RDSGBLDEC 302 couples the segmentbitlines to the top bitlines when selected. The RDSGINHDEC 303 couplesthe inhibit voltage (VINH) 999 to the appropriate bitlines of theselected array units (ARYSEG) 290 when selected or unselected asdescribed later in FIG. 9B. The RD1SUBBLK 304 enables appropriatecontrol gates and common lines for the memory cells.

FIG. 8 shows details of the power supply decoder (RDSGPSDEC) 301. Line(NI) 310 and (OI) 311 are predecoded address lines coming from theaddress predecoder block (XPREDEC) 154. Line ENVSUPDEC 312 is a globalenable signal for disabling or enabling all the supply decoders. A NANDgate 315 is a typical 3-input NAND gate with an output line (ENB) 313.An inverter 316 is a typical inverter with input line (ENB) 313 and anoutput line 314. A high voltage level shifter (HVLS1) 317 shifts logicsignal EN 314 into high voltage complementary output signal lines(ENVSUPB) 318 and (ENVSUP) 319. A line (VXRGND) 333 is a low voltageline for (HVLS1) 317. A line (VHSUPPLY) 777 is a precisely regulatedhigh voltage supply for the decoding. A line (VMSUPPLY) 666 is anotherprecisely regulated high voltage supply. A transistor PMOS 322 couplesthe high voltage supply (VHSUPPLY) 777 into line (VHSUPPLYSG) 328 whenthe RDSGPSDEC 301 is selected. Transistors PMOS 323 and 324 coupleregular voltage supply (VDD) 1111 into line (VHSUPPLYSG) 328 when theRDSGPSDEC 301 is deselected. A transistor PMOS 325 couples another highvoltage supply (VMSUPPLY) 666 into line (VMSUPPLYSG) 329 when theRDSGPSDEC 301 is selected. The voltage level on line (VMSUPPLY) 666,e.g., 5-10V, is such that in read the bitline select transistors in thememory array are heavily overdriven to reduce their on resistance, whichresults in insignificant sense error. Transistors PMOS 326 and 327couple regular voltage supply (VDD) 1111 into line (VMSUPPLYSG) 329 whenthe RDSGPSDEC 301 is deselected. The PMOS 323 and 326 have their wellsconnected to line (VDD) 1111. The PMOS 324 and 327 have their wellsconnected to the VHSUPPLYSG 328 and VMSUPPLYSG 329, respectively. Theconnection of their wells is done to avoid source and drain junctiondiodes turning on during the switching.

FIG. 9A shows details of the segmented bitline select decoder(RDSGBLDEC) 302. Line (ENVSUP) 319 and line (ENBLAVH) 341 connected tothe gates of transistors 360 and 361, respectively, are used to couplevoltage on line VMSUPPLYSG 329 into line ENBLA 369. Either transistor362 with line (ENB) 313 on its gate or transistor 363 with line(ENBLBVL) 342 on its gate is used to couple line (ENBLA) 369 to line(VXRGND) 333. Similarly transistors 364 and 365 together with lines(ENVSUP) 319 and line (ENBLBVH) 343, respectively, on their gates areused to couple voltage on line (VMSUPPLYSG) 329 into line (ENBLB) 371.Either transistor 366 with line (ENB) 313 on its gate or transistor 367with line ENBLAVL 340 on its gate are used to couple line (ENBLB) 371 toline (VXRGND) 333. The voltage level on line (VHSUPPLY) 777 in the block(RDSGPSDEC) 301, e.g., 7-12 V, is such that the transistors 360, 361,364, 365 couple, with minimal loss, the voltage from VMSUPPLYSG 329 intolines (ENBLA) 369 and (ENBLB) 371. The deselect transistors 362, 363,366, and 367 have their gates coupled only to the low voltage signalsinstead of the high voltage control signals as conventionally done. Thiscircuit configuration has the benefit of reducing significantly theloading for the high voltage supply (VHSUPPLY) 777. This circuitconfiguration is applied throughout all the decoding circuits.

FIG. 9B shows details of the segmented inhibit select decoder(RDSGINHDEC) 303. Either transistor 350 with line (ENVSUPB) 318 on itsgate or transistor 353 with line (ENBLBVH) 343 on its gate couples thevoltage on line (VMSUPPLYSG) 329 to line (INHBLA) 345. Transistors 351and 352 together with lines (EN) 314 and (ENBLAVL) 340, respectively, ontheir gates are used to couple line (INHBLA) 345 to line (VXRGND) 333.Similarly either transistor 354 with line (ENVSUPB) 318 on its gate ortransistor 357 with line (ENBLAVH) 341 on its gate is used to couple thevoltage on line (VMSUPPLYSG) 329 to line (INHBLB) 347. Transistors 355and 356 together with lines (EN) 314 and line (ENBLBVL) 342 respectivelyon their gates are used to couple line (INHBLB) 347 to line (VXRGND)333. Transistor 358 with line (ENVSUP) 319 on its gate is used to couplethe inhibit voltage on line (VINH) 999 to line (VINHSEG) 349. Transistor359 with line (ENB) 313 on its gate is used to couple the voltage online (VINHSEG) 349 to line (VXRGND) 333. Similar to the circuitconfiguration in the RDSGBLDEC 302, the low voltage signals are used forsignal deselection.

The circuit blocks RDSGPSDEC 301, RDSGBLDEC 302, RDSGINHDEC 303, andRD1SUBBLK 304 are used in the array as shown in FIG. 4A for arrayselection and inhibit decoding.

FIG. 9C shows a predecoded common line segmented decoder (RDSGCLPDEC)302B for lines (CLP0-3) 445A-D. Lines (CLP0-3) 445A-D come from a commonline pre-decoder (XCLPREDRV) 950. The purpose of this circuit(RDSGCLPDEC) 302B is to greatly reduce the capacitive loading on linesCLP0-3 seen by the common line pre-decoder (XCLPREDRV) 950. Lines(CLPS0-3) 456A-D are the output lines. Transistors 438A-D with line(ENVSUP) 319 on their gates are used to couple lines (CLP0-3) 445A-D tolines (CLPS0-3) 456A-D, respectively. Transistors 439A-D with line (ENB)313 on their gates are used to couple lines (CLPS0-3) 456A-D to line(VXCLGND) 5555. This concept of segmented loading could also be appliedto predecoded control gates CGP0-15.

FIG. 10 shows details of the sub-block decoder (RD1SUBLK) 304, thatincludes a circuit block 304A and a circuit block 304B. The block 304Aincludes a NAND gate 412, an inverter 413, and a high voltage levelshifter (HVLSX) 418. The 3-input NAND gate 412 is used for addressdecoding. Line (ENB4) 414 is its output. Lines (MI) 410, (NI) 310, and(OI) 311 are predecoded address lines coming from the addresspre-decoder (XPREDEC) 154. The inverter 413 inverts line (ENB4) 414 intoline (EN4) 415. The high voltage level shift (HVLSX) 418 is used toshift the logic signal EN4 415 into the high voltage output signal(ENHV4BLK) 417. Line (VHSUP) 770 supplies high voltage for the levelshifter (HVLSX) 418. Line (VHSUP) 770 couples to line (VHSUPLYSG) 328 ofcircuit block (RDSGPSDEC) 301. The circuit block 304B including a set offour circuit blocks (RD4CG1CL) 416 provides control signals for controlgates (CG) and common lines (CL). Lines CG[0:15] 422A-P couple to 16rows of memory cells, for example, lines 262, 263, 265-268, 270, 271 ofthe block (ARY1BLK) 290A in FIG. 4A. Lines CL[0:3] 423A-D couple to 4shared common lines of memory cells, for example, lines 264 and 269 ofthe block ARY1BLK 290A in FIG. 4A. Lines CGP[0:15] 420A-P are predecodedcontrol gate lines coming from the control gate pre-decoder (XCGPREDEC)951. Lines CLPS[0:3] 456A-D are predecoded common lines coming fromblock RDSGCLPDEC 302B. Line (VXCGGND) 444 is a line for control gate(CG) deselection. Line (VXCLGND) 5555 is a line for common line (CL)deselection.

FIG. 11A shows details of circuit block (RD4CG1CL) 416. Transistors 430,432, 434, 436 together with lines (CGP0) 440, line (CGP1) 441, line(CGP2) 442, line (CGP3) 443, respectively, on their drains are used tocouple these lines 440-443 to output line (CG0) 450, line (CG1) 451,line (CG2) 452, and line (CG3) 453, respectively. Lines (CGP0-CGP3)440-443 come from a control gate predecoder (XCGPREDEC) 951. Transistor438 is used to couple line (CLPS0) 456A to line (CL0) 454. Transistor439 is used to couple line (CL0) 454 to line (VXCLGND) 5555. Line(ENHVLBLK) 446 couples high voltage into the gates of transistors 430,432, 434, and 436. Line (ENB1BLK) 447 couples lines (CG0-3) 450-453 tothe line (VXCGGND) 444 through transistors 431, 433, 435, and 437,respectively, and couples line (CL0) 454 to line (VXCLGND) 5555 throughtransistor 439. The lines (ENHV1BLK) 446 and (ENB1BLK) 447 are coupledrespectively to lines (ENHV4BLK) 417 and (ENB4) 414 generated by circuitblock 304.

Four common lines of memory cells are coupled together to one decodedcommon line CL as shown in the block (ARYSEG0) 290 in FIG. 4A. Fourblocks of the RD4CG1CL 416 are used to provide array block selection asshown in the block (ARYSEG0) 290 in FIG. 10. One array block is definedas including 16 rows and 4 common lines of memory cells. One array blockincludes a plurality of blocks (ARY1BLK) 290A connected horizontally.

The lines (VXRGND) 333, (VXCLGND) 5555, and (VXCGGND) 444 could beindividually controlled to be biased at different voltage levels duringerase, read, and program to optimize circuit functionality, forinstance, to increase the breakdown or to reduce the leakage of MOSdecoding transistors.

Note that the same transistors are used for decoding in erase, read, andprogram operation. In conventional decoding, read decoding is isolatedfrom erase and program decoding since read decoding requires only lowvoltage and hence the decoding size can be optimized for read speed.Here all decoding is combined together to minimize the die size. Furtherall decoding is done by NMOS transistors instead of by both PMOS andNMOS transistors as conventionally done. This has the benefit ofreducing the capacitive loading. This is so because in deselection onePMOS presents itself as a gate capacitor load while one NMOS onlypresents itself as a source or drain overlap capacitor load, which ismuch smaller than a gate capacitor load. Low capacitive loading leads toless power consumption for NMOS decoding. This is against conventionalwisdom, which holds that a CMOS circuit is more power efficient than aNMOS circuit.

FIG. 11B shows an alternative circuit block (RD4CG1CL) 416 with adiode-connected transistor 438F. The transistor 438F provides feedbacksignal (CLK) 445F for a Kelvin type connection to a circuit driverinside the block (XCLPREDRV) 950. A Kelvin connection line consumesminimal (or no) DC current. A Kelvin connection allows a circuit driversuch as a common line circuit driver to stabilize its output signal at adesired voltage level based on feedback voltage from the Kelvinconnection line. This Kelvin connection line (CLK) 445F is connected toother Kelvin connection lines vertically. This is possible since onlyone common line is on at any given time. Once a common line is selected,this common line will take control of the CLK 445F line since thediode-connected transistor will be forward biased and otherdiode-connected transistors on the rest of the common lines will bereverse biased. This will be known as winner-take-all Kelvin decoder.This winner-take-all Kelvin decoder will ensure a predetermined voltageon the line (CL0) 454 will be stable all the time over varying load,process corners, temperature, and power supply variation with minimumcost. The stable voltage on the common line is required to not introducesignificant voltage error in program or in read as described previouslyin the description of the multilevel array architecture.

FIG. 11C shows a circuit block (RD1CL) 304C, which is used in a commonline segmentation scheme with the array partitioning shown in FIG. 12 toreduce the voltage drop along the common lines. In an embodiment, onecommon line (CL) is connected together across the full array with aplurality of blocks (RD1CL) 304C driving the same common line (CL).Transistor 438S with line (ENHV1BLK) 446 on its gate couples line(CLPS0S) 456AS to line (CL0) 454. Line (CL0) 454 of this circuit block304C is the same line (CL0) 454 of the circuit block (RD4CG1CL) 416. Adeselect transistor 439S with line (ENB1BLK) 447 couples line (CL0) 454to line (VXCLGND) 5555. The transistor 439S is optional in this circuitsince the function of coupling line (CL0) 454 to line (VXCLGND) 5555 isalready provided by the transistor 439 in the RD4CG1CL 416. Thetransistor 439S provides additional drive ability in addition to that ofthe transistor 439. Line (CLPS0S) 456AS couples to a common linepre-decoder (XCLPREDRV) 950. The winner-take-all Kelvin decoding canalso be used here. The control signals (ENHV4BLK) 417 and (ENB4) 414shown in the block (RD1SUBBLK) 304 couple to control signals (ENHVLBLK)446 and (ENB1BLK) 447, respectively. The control signals (ENHV4BLK) 417and (ENB4) 414 are fed through the memory array as shown in FIG. 12. Inan alternate embodiment, one common line is divided into many separatecommon lines across the full array. These separate common lines are notconnected to each other. In this case, each separate common line isdriven on both sides by two blocks (RD1CL) 304C or by a (RD1CL) 304C anda (RD4CG1CL) 416. Common line segmentation is described more in detailbelow in description associated with FIG. 12.

FIG. 12 shows a feedthrough-to-memory and feedthrough-to-driver schemetogether with the common line segmentation to deliver precise voltagesfor memory cells as described in the following. The feedthrough schemeexploits the multi-layer metal interconnect to reduce the circuitcomplexity and die size and to enable innovative circuit configurations.A conventional flash memory system typically only uses up to a maximumof 2 metal layers and hence is limited in core interconnect schemepossibilities. This feedthrough scheme is made possible by three or moremetal layers.

The block (MLMDECS) 132, shown in FIG. 12 and also in FIG. 3A, includesa plurality of the blocks (RDSGCLPDEC) 302B and a plurality of theblocks (RD1CL) 304C. Only one block (RDSGCLPDEC) 302B and one block(RD1CL) 304C per block 132 are shown in FIG. 12 for clarity. Otherblocks have similar connections. The block (MLMDEC) 130, shown in FIG.12 and also in FIG. 3A, includes a plurality of the blocks (RD1SEG) 300.The block RD1SEG 300 includes a block (RDSGPSDEC) 301 and a plurality ofblocks (RD1SUBBLK) 304. Only the block (RDSGPSDEC) 301 and one block(RD1SUBBLK) 304 inside one block RDLSEG 300 are shown in FIG. 12 forclarity. Other blocks have similar connections.

The feedthrough-to-memory uses a single driver to drive both left andright sides of a memory array. The layout of row decoding circuits suchas of the block (RD1SUBBLK) 304 is very dense because of the limitedheight of a typical advanced memory cell, e.g., 0.5-1 μm per cellheight, and the very wide width of each decoding transistor, e.g., 20-50μm, due to their required precision multilevel drive ability. This makesit extremely difficult to route the required lines from the right sideacross the active circuit of this row decoding circuit to the left sidewith limited layers of metal interconnect. As shown in FIG. 10, thecontrol lines CG[0:15] 422A-P and common lines CL [0:3] 423A-D providesthe control signals to the memory cells on the right side as well as thememory cells on the left side. This is also shown in FIG. 12 in block304B with lines pointing to the right as well as to the left. Similarlyit also shows the control lines from circuit block 304A and 304C drivingboth sides. The feedthrough-to-memory scheme also shows predecoded highvoltage lines (ENHV4BLK) 417 and (ENVSUP) 319 and predecoded low voltagelines (ENB) 313 and (ENB4) 414 being fed through the memory by runningon top of the memory, for example, in metal 4, without interfering withthe memory cells underneath. Other control lines could also be fedthrough the memory. Again this is achievable by three or more metallayers which allow a different circuit configuration with minimal activearea. The circuit block 304C is the precision voltage driver for thecommon lines CL of the memory cells in addition to the circuit block304B. The feedthrough-to-driver scheme shows control signals fromcircuit blocks 304B and 304A being fed through the memory array to theprecision voltage drivers 304C.

The common line segmentation is also shown in FIG. 12. Each metal commonline runs the length of the memory core horizontally across the fullarray with seven circuit blocks (RD1CL) 304C and two circuit blocks(RD1SUBBLK) 304 driving the same common line. The voltage drop acrossone common line is thus divided into eight voltage drop segments. Eachvoltage drop segment belongs to each common line of each sub-array block(MFLSUBARY) 101. Within each voltage drop segment, the voltage value onthe left side is same as the voltage value on the right side of thevoltage drop segment and the lowest voltage value is in the middle ofthe voltage drop segment. This is because there is a precision circuitdriver (RDLCL) 304C or (RD4CG1CL) 416 on each side of the voltage dropsegment. One alternative embodiment of the common line segmentationscheme is to have these common lines physically divided into eightseparate common lines. That is, each sub-array block (MFLSUBARY) 101shown in FIG. 12 has its separate common line. However, in this case,the deselect transistor 439S in the block (RD1CL) 304C is no longeroptional but necessary to deselect each separated common line.

The voltage level on the control gates is controlled by the voltage onthe lines (CGP[0:15]) 420A-P in circuit block 304. The voltage on lines(CGP[0:15]) 420A-P are in turn controlled by a precise bandgap-referredregulated voltage. Hence precision voltage level is provided at thememory control gates. The voltage level on the common lines iscontrolled by the voltage on the predecoded common lines (CLP[0:3])421A-D in circuit block 304. The voltage on lines (CLP[0:3]) 421A-D arein turn controlled by a precise bandgap-referred regulated voltage foreach common line driver. Hence precision voltage level is provided atthe memory common lines. The programming and sensing current bias arealso bandgap-referred; hence they are highly stable.

Note that in FIG. 12 an alternative embodiment is to share one block(RDSGPSDEC) 301 or 304A across the full array by doing feedthrough ofthe outputs of (RDSGPSDEC) 301 or 304A across the full memory array. Inthis case the drive ability of the driver circuit inside block(RDSGPSDEC) 301 or 304A should be adequately designed to drive the longinterconnect lines.

Note that in FIG. 10 an alternative embodiment is to have a separateblock (RD4CG1CL) 416 for driving the right side of an array and anotherseparate block (RD4CG1CL) 416 for driving the left side of an array.Another alternative embodiment is to share just one CL driver for bothleft and right sides but to have separate control gate CG drivers forthe left side and the right side.

Multilevel Reference System:

FIG. 13 shows a block diagram for a multilevel digital memory referencesystem. All the relevant blocks have been described in association withprevious figures. The highlighted blocks 106, 116, 126, and 146 with thehighlighted lines (VREF0-15) 760-775 are shown to show the referencesystem in relation to the physical position of the array and y-drivers.The physical position of the reference array corresponding to variousschemes is explained in the following description.

FIG. 14 shows details of a multilevel digital memory reference system. Areference circuit block (VREFGEN) 719 is used to provide all referencevoltage levels for erasing, programming, sensing, margin tests, andproduction tests. Shown are reference levels for reference cells(VREFR0-15) 700-715 and reference levels for data cells (VREFD0-15)720-735. Data cells refer to memory cells that store digital data. A 16level multilevel flash cell is assumed for this discussion. A flashreference array (MFLASHREF) 106 includes a plurality of blocks(MFLASHREFS) 106A. A block (MFLASHREFS) 106A includes a plurality ofreference memory cells. A reference page select 126A is used to selectthe reference cells in the blocks (MFLASHREFS) 106A associated with aselected page. Each block 126A selects one reference cell in onecorresponding block (MFLASHREFS) 106A. For each selected page, there are16 blocks 126A selecting 16 reference cells in 16 corresponding blocks(MFLASHREFS) 106A. The 16 selected reference cells makes up one pagereference.

A buffer (VRBUFFER) 750 and a comparator 801 are inside a block(REFYDRVS) 116S. The buffer (VRBUFFER) 750 is used to drive eachreference level of (VREF0-15) 760-775 for all the y-drivers. A buffercircuit without offset auto zero 750A is used to isolate the referencecell from all capacitance from auxiliary circuits. The offset auto zerocancels out the voltage offset of an analog buffer. The voltage offsetof an analog buffer is typically uncontrollable and is caused bythreshold voltage mismatch, transistor transconductance mismatch, andsystematic offset. This voltage offset would cause an uncertainty in thereference voltage, which would degrade the margin of one voltage levelwith respect to another voltage level. Line (VBUFO) 781 is used toverify a reference cell is programmed to one desired reference level outof 16 possible reference levels. Line (VBUFO) 781 is used instead of thedirect memory cell output for verifying in the verify cycle. This is toinclude the buffer offset from buffer 750A in the verifying process. Thecomparator 801 is used to do the actual comparison in verify. A bufferwith offset auto zero 750B is used to drive a reference level. Variousvoltage levels needed for multilevel algorithm are also generated by thebuffer 750B with switch capacitor technique. The auto zero is needed tozero out the offset of this buffer since a typical buffer offset is10-20 mV. This voltage amount if not canceled out would degrade themargin of a reference level, which effectively reduces the voltagemargin for each level. Capacitors are needed to accomplish the auto zeroand level shifting operation in the buffer 750B. However as described inthe array architecture description, any additional capacitance wouldadversely degrade the write and read speed. Hence buffer 750A isinserted between the reference cell and the buffer 750B so that thereference cell only sees one gate capacitance inside a typical buffer asa capacitor load.

Lines (VREF0-15) 760-775 are the final reference lines driving into allthe y-drivers as needed for verify-program cycles and read cycles.Switch S 750D couples line (VREFD) 720 to the input terminal of buffer750B when one selected page programs for the first time. Switch S 750Ccouples line (VBUFO) 781 to input terminal of buffer 750B when the sameselected page programs for the second time or more without an erase inbetween program. The reason is that for first time programming,reference levels for data cells come from a reference generator VREFGEN719 and for subsequent programming reference levels come from thereference cells in MFLASHREFS 106A.

For the memory system described herein, there are 8 pages for each row,4 rows for each block, and 512 bytes per page with a 4-bit digitalmultilevel memory cell. Since any one page is written or read at anytime a complete reference set of 16 levels is reserved for each pageinstead of for each row. This is done to preserve the operatingconditions through the lifetime of a memory system exactly the same forreference cells as regular data cells. This is done for example to makethe reference and data cells have the same voltage readout drift overtime. For each row, there are 8×16=128 reference cells. This has somesmall die size penalty. The reference cells are written at the same timeas the regular data cells.

After the reference cells are written with the first programmingsequence, if subsequent programming cycles are allowed to write otherdata cells in the same page, the previously programmed reference cellsstay in the program inhibit mode. This is accomplished as shown in FIG.15. A comparator 850 is used to compare a reference voltage from abandgap VREF 851, e.g., 1.2 V, versus a readout voltage from a referencememory cell VREFOUT 852, for example, level 0, e.g., 0.5V. If thereference cell has not been written, VREF 851<VREFOUT 852, then line(REFON) 853 would be low. If the reference cell has been written, VREF851>VREFOUT 852, then line (REFON) 853 would be high indicating that thereference cells have been previously written and the reference cells areinhibited in programming.

For subsequent programming cycles after the first programming cycle, thereference voltages for the data cells come from the reference cells andthe reference voltages are shifted appropriately to place the datavoltages in between the adjacent reference voltages.

The voltage drop along the common line poses a particular problem for amultilevel reference system. Reference cells are needed to track thedata cells over temperature, process, or power supply. But astemperature changes, the voltage drop along the common line changes,which causes a sense error. The voltage drop along the line from one endto the other end follows geometrically as described earlier. That isdepending on position along the common line, the cells experiencedifferent amounts of common line voltage changes, which cause differentvoltage readout shifts due to different voltage amounts being coupledinto the cells. This cannot be corrected by a conventional referencesystem.

FIG. 16 shows a positional linear reference system that corrects thiserror. Assuming the voltage drop along a line is linear and assuming anacceptable voltage shift is DVREF/2, by dividing the voltage dropDVTOTAL 859=VBEG 855−VEND 856, into different voltage segments withequal voltage drop DVREF 858 and by positioning the reference cells 857in the middle of a divided array segment (ARYVSUB1-3) 888A-Ccorresponding to a voltage segment, the maximum voltage difference froma reference cell to a data cell in the beginning or at the end of thevoltage segment is =<DVREF/2. Hence reference correction overtemperature is achieved. It is possible to place the reference cells 857at the beginning or the end of a divided array segment (ARYVSUB1-3)888A-C. In this case the maximum voltage difference from a referencecell to a data cell is DVREF instead of DVREF/2 as in the case ofpositioning the reference array in middle of a divided segment array.Another advantage of placing the reference cells in the middle of adivided array segment is to minimize the electrical variation due to theedge interface from the memory array to peripheral circuits.

FIG. 17 shows a positional reference geometric system basing on theconcepts similar to FIG. 16. In this embodiment, the reference cells 857are not symmetrically but geometrically positioned to correct for thegeometric effect of the voltage drop.

In FIGS. 16 and 17, each full array is divided into three sub-arrays(ARYVSUB1-3) 888A-C and (ARYVSUB4-6) 888D-F respectively. It should benoted that the array could be divided into as many sub-arrays as neededto reduce the voltage error. Also shown in FIGS. 16 and 17, eachsub-array of ARYVSUB1-6 888A-F includes its own complete set ofreference cells in the middle. A complete set of reference cellsprovides all the reference levels, e.g., 16 levels for 4-bit digitalmultilevel cell per page, for all the pages. One row of reference cellsincludes 128 reference cells if each row has 8 pages and each referencecell provides one reference level. An alternative embodiment is to havemore than one reference cell per level, e.g., 4-16 cells per level. Thisaverages out the electrical variation of multiple cells.

FIG. 18 shows a geometric compensation reference system. The objectiveis to simulate the voltage drop in the common line into the referencereadout voltage by attaching similar loading currents to the referencereadout voltage. A resistance R 862 in the reference line is madeequivalent to a resistance R 866 in the common line. A reference loadingcurrent (ICELLR) 868R is made the same as that of ICELL 868. Hence thetotal voltage drop in reference DVREFTOTAL 863, =REFB 860−REFE 861, isequal to DVCLTOTAL 867, =VCLB 864−VCLE 865. It is not necessary toattach the same number of loading reference currents ICELLR 868R to thenumber of ICELL 868. It is only necessary to attach the approximateamount of the current loading at appropriate positions to minimize theerror to an acceptable level.

One alternative embodiment of the reference system is, instead of using16 reference cells for a 4-bit digital multilevel cell, to use 2 or 4 or8 reference cells to generate 16 reference levels with levelinterpolation. That is from reference levels coming from referencecells, the other reference levels are interpolated by using linear orany other interpolation.

Multilevel Algorithm:

FIG. 19A shows various voltages generated and used in one embodiment ofthe invention for program verifying, program upper and lower marginverifying, read sensing and restore high or restore low margin verifyingduring read sensing. The read sensing is advantageously performed in thevoltage-mode but other modes of read sensing are also applicable. Allthe voltages are generated by the V&IREF block 172. VREFR(L) is theprogram verify voltage used to verify program level (L) of a referencecell. VREFD(L) is the program verify voltage used to verify programlevel (L) of a data cell. For example, in a 4 bit per cell storageembodiment there are 16 levels used. It is also possible to use 15levels instead of 16 levels since the extreme low or high levels notneed to be constrained to exact low or high levels but can go to groundor power supply respectively. VREFR0 through VREFR15 are program verifyvoltages used for verifying programming of the reference cells. VREFD0through VREFD 15 are program verify voltages used for verifyingprogramming of the data cells. VUM(L) and VLM(L) are upper and lowerprogram margin voltages respectively for level L. Each level L may haveits own VUM(L) and VLM(L) voltage values. VUM(L) and VLM(L) can each beof different value also for each level L. On the other hand, VUM(L) andVLM(L) can be of the same voltage value for all the levels. VUM(L) andVLM(L) voltages are generated by the block V&IREF 172. VRSTH(L) andVRSTL(L) are RESTORE HIGH and RESTORE LOW margin voltages respectivelyfor level L. Each level L may have its own VRSTH(L) and VRSTL(L) voltagevalue. VRSTH(L) and VRSTL(L) can each be of different value also foreach level L. On the other hand, VRSTH(L) and VRSTL(L) can be of thesame voltage value for all the levels. VRSTH(L) and VRSTL(L) voltagesare generated by the V&IREF 172 block. VCELLR(L) is the voltage readback from a reference cell during read sensing. VCELLD(L) is the voltageread back from a data cell during read sensing. The cross-hatchedregions show the distribution of possible read back voltages during readsensing after reference cells or data cells have been programmed to acertain level L, while using VREFR(L) or VREFD(L) as the program verifyvoltage, respectively. The distributions occur because every cell doesnot have the same programming or read sensing characteristics.

Page Programming Cycle:

FIG. 20 shows the flow diagram for one embodiment of the pageprogramming cycle. During a page programming cycle a plurality of memorycells are programmed in parallel. However this algorithm is equallyapplicable for single cell programming. As an example, 4 bit per cell isprogrammed in each cell. First the program command is issued and theaddress of the page to be programmed is loaded. The data count NC isinitialized. The address loading may be performed through a single or aplurality of address cycles. Program data is input during the DATAINstep and is selectively loaded in the internal latches of a YDRVS 110Sor SYDRVS 114S or RYDRV 112S. Block YDRV 110, SYDRV 114, (RYDRV) 112includes a plurality of YDRVS 110S, SYDRVS 114S, RYDRVS 112Srespectively. Block YDRVS 110S will be described in detail later in thedescription associated with FIG. 26. Data gets loaded into the datalatches of the current YDRVS 110S or SYDRVS 114S selected from theADDRCTR 162 and the BYTEDEC 152. The redundancy control block REDCNTRL186 asserts RED_ADD_TRUE true (YES or Y) or false (NO or N) to signifywhether the current YDRVS 110S or SYDRVS 114S is GOOD or BAD. A YDRVS110S or SYDRVS 114S is GOOD if it has not been flagged as one thatcannot be used to load input data on its data latches. A YDRVS 110S orSYDRVS 114S is BAD if it has been flagged as one that cannot be used toload input data on its data latches. GOOD or BAD YDRVSs or SYDRVSs areflagged during manufacturing testing and the flags are internally storedon non-volatile latches. If RED_ADD_TRUE=NO, meaning current YDRVS 110Sor SYDRVS 114S is GOOD, then a data nibble on the IO[0:3] or IO[4:7] busis placed at the input of the data latches of the current YDRVS 110S orSYDRVS 114S. A data byte consists of 8 digital bits and a data nibbleconsists of 4 digital bits. If RED_ADD_TRUE=Y, meaning current YDRVS110S or SYDRVS 114S is BAD, then the data nibble on the IO[0:3] orIO[4:7] bus is placed at the data latches of the selected RYDRVS 112S.Next, if NEXTDATAIN=Y, the data at the input of the data latches of therespective YDRVS 110S, SYDRVS 114S or RDYRVS 112S is latched. IfNEXTDATAIN=N then the flow waits for the program start command PRG.Next, if the data count NC>MAXNC=not true (N), then NC=NC+1 and the flowloops back to DATAIN step to load in the next data byte. If the datacount NC>MAXNC=true (Y), then the flow goes out of the loop and waitsfor the program start command PRG. The data count MAXNC signifies thenumber of data bytes that are simultaneously programmed in a page. Next,if command PRG is received then page programming begins. If command PRGis not received then the flow loops back to check for NEXTDATAIN. Nodata loading is required for blocks (REFYDRVS) 116S because theirlatches are internally set. A block (REFDRV) 116 includes a plurality ofblocks (REFYDRVS) 116S.

FIG. 21 shows the flow diagram after page programming begins. TheProgram flag=Pass is set and the BUSY signal is set. In anotherembodiment a configuration (fuse) bit initialization is executed to loadin data from fuse non-volatile memory cells to volatile latches locatedin the fuse circuit block (FUSECKT) 182 at this step. The programinhibit mode of all cells in the page being programmed are reset toenable programming. Based on the output B[0:3] of the data latches ofeach YDRVS 110S, SYDRVS 114S or RYDRVS 112S a program verify voltageVREFD(L) is set at the input of the comparator in each of the respectiveYDRVS 110S, SYDRVS 114S or RYDRVS 112S. Based on the output B[0:3] ofthe data latches of each REFYDRVS 116S a program verify voltage VREFR(L)is set at the input of the comparator in each REFYDRVS 116S. For eachreference cell and data cell in the page being programmed, the cellvoltage VCELLD(L) or VCELLR(L) is read. Depending on the output B[0:3]of the data latches (a) for each REFYDRVS 116S the appropriate programverify voltage VREFR(L) is compared to the reference cell read backvoltage VCELLR(L) and (b) for each YDRVS 110S, SYDRVS 114S, RYDRVS 112S,the appropriate program verify voltage VREFD(L) is compared with datacell read back voltage VCELLD(L) to indicate whether further programmingis required. If no further programming is required for a particularreference cell or data cell, it is put in the program inhibit mode. Ifthe Program Pulse Count=MAXPC is not true, then the cells are placed inthe program mode and another programming pulse is applied to all thecells in the page, including the reference cells. Cells which are in theprogram inhibit mode do not get any additional programming. Cells whichare not in the program inhibit mode get additional programming. Afterthe programming pulse is applied, the program pulse count is incrementedand the cells are placed in the voltage-mode read to verify if furtherprogramming is required. This iterative verify-program loop is continueduntil either all the cells in the page including the reference cells arein the program inhibit mode or when the program pulse count=MAXPC istrue. If program pulse count=MAXPC true condition is reached, before allcells in the page including the reference cells are all in programinhibit mode, then the program flag=fail condition is set, BUSY signalis reset and the programming cycle is done. Whenever the All Cells inProgram Inhibit Mode=true condition is reached, the flow moves to thenext step as shown in FIG. 22A.

As shown in FIG. 22A, next, for each level L, upper program marginverify voltage UMV(L)=VCELLR(L)−VUM(L) is generated, where VUM(L) is theupper margin voltage for level L. Depending on the data latch outputB[0:3] of the data latches in the respective YDRVS 110S, SYDRVS 114S,RYDRVS 112S the appropriate voltage UMV(L) is compared with read backcell voltage VCELLD(L) for all the data cells. If the result ofcomparison indicates that all upper cell margins are not within limitsthen a program flag=fail condition is set; BUSY signal is reset andprogramming cycle is done. If the result of comparison indicates thatall the upper cell margins are within limits then a program flag=failcondition is not set and then, for each level L, lower program marginverify voltage LMV(L)=VCELLR(L−1)+VLM(L) is generated, where VLM(L) isthe lower margin voltage for level L. Depending on the data latch outputB[0:3] of the data latches in the respective YDRVS 110S, SYDRVS 114S,RYDRVS 112S the appropriate voltage LMV(L) is compared with read backcell voltage VCELLD(L). If the result of comparison indicates that alllower cell margins are not within limits then a program flag=failcondition is set; BUSY signal is reset and programming cycle is done. Ifthe result of comparison indicates that all the lower cell margins arewithin limits then a program flag=fail condition is not set and BUSYsignal is reset and programming cycle is done. The program flag=failindicates the programming cycle has been unsuccessful to program thecurrent page. It does not indicate specifically which cell or cellscaused the unsuccessful programming.

Page Read Cycle:

FIG. 23 shows the flow diagram for the page read cycle. During a pageread cycle a plurality of memory cells are read in parallel. Howeverthis algorithm is equally applicable for single cell read. After thepage read command is issued along with the address of the page to beread, the BUSY signal is set, RESTOREL and RESTOREH flags are reset, thedata latches in the YDRVS 110S, SYDRVS 114S, RYDRVS 112S are set tooutput B[0:3]=1111 and N is set to 3. N represents the number of bitsstored per memory cell. In another embodiment a configuration (fuse) bitinitialization is executed to load in data from fuse non-volatile memorycells to volatile latches located in the fuse circuit block (FUSECKT)182 at this step. All the cells in the addressed page are placed in thevoltage-mode read and the cell voltages, VCELLR(L) for reference cellsand VCELLD(L) for data cells are read. BN is forced to “0” and the readverify voltage VCELLR(L), which is one of the reference read backvoltages dependent on B3, B2, B1, B0, is compared with the cell readback voltage VCELLD(L). For each cell, if the VCELLD(L)>VCELLR(L) thenBN is latched as “1”, otherwise BN is latched as “0”. The loop continuesuntil all the bits B3, B2, BE1, B0 are latched and N=0. Next, as shownin FIG. 24, for each level L, a MARGIN RESTORE LOW VoltageVRSTRL(L)=VCELLR(L)−VRSTL(L) is generated, where VRSTL(L) is the restorelow margin voltage. Depending on the latched bits B3, B2, B1, B0 on eachof the YDRVS 110S, SYDRVS 114S, RYDRVS 112S, the voltage VRSTRL(L) iscompared with the respective data cell read back voltage VCELLD(L). IfVCELLD(L)>VRSTRL(L) for any one of the cells, then the RESTOREL flag isset. Next, for each level L a MARGIN RESTORE HIGH VoltageVRSTRH(L)=VCELLR(L−1)+VRSTH(L) is generated, where VRSTH(L) is therestore high margin voltage. Depending on the latched bits B3, B2, B1,B0 on each of the YDRVS 110S, SYDRVS 114S, RYDRVS 112S, the voltageVRSTRH(L) is compared with the respective data cell read back voltageVCELLD(L). If VCELLD(L)<VRSTRH(L) for any one of the cells, then theRESTOREH flag is set, otherwise RESTOREH flag is not set. Next, as shownin FIG. 25, BUSY signal is reset and the byte count ND is initialized toNDI. NDI is the byte count of the existing byte address location. Allbits in the respective YDRVSs, SYDRVSs, or RYDRVSs data latches areready to be sequentially read. Whenever READ CL0CK=Y, the RED_ADD_TRUEis checked for that byte address location. If RED_ADD_TRUE=Y, then datafrom RYDRVS 112S is output to the IO port IO[0:7] 1001, otherwise datafrom YDRVS 110S is output to the io port IO[0:7] 1001. If READ CL0CK=Nand ENABLE=Y then the flow loops back until READ CL0CK=Y or ENABLE=N.After all the data is output i.e. ND>MAXND=Y or if ENABLE=N, the Pageread cycle is done. If ND>MAXND is =N, then ND is incremented and theflow loops back to check the READ CL0CK.

FIG. 26 shows the details of an embodiment of YDRVS 110S. SYDRVS 114Sand RYDRVS 112S have similar details. The description given for YDRVS110S is equally applicable for SYDRVS 114S and RYDRVS 112S. In thisembodiment 4 bits are stored per memory cell, hence four data latchesare required per YDRVS 110S. A set of four data latches (DATALAT3) 10,(DATALAT2) 11, (DATALAT1) 12, (DATALAT0) 13 holds the data during theDATAIN step of a page programming cycle or holds the data during a LATCHEN=1 or =0 step during a page read cycle. Data is loaded into DATALAT310, DATALAT2 11, DATALAT1 12, DATALAT0 13 through the DIN3 14, DIN2 15,DIN1 16, DIN0 17 lines respectively and read out from the DATALAT3 10,DATALAT2 11, DATALAT1 12, DATALAT0 13 through the DOUT3 18, DOUT2 19,DOUT1 20, DOUT0 21 lines respectively. Lines (DIN3) 14, (DIN2) 15,(DIN1) 16, (DIN0) 17, (DOUT3) 18, (DOUT2) 19, (DOUT1) 20, (DOUT0) 21connect to BYTESEL 140 for YDRV 110 and connect to blocks 144, 142 forSYDRV 114, RDYRV 112 respectively. During page program cycle, lines (B3)22, (B2) 23, (B1) 24, (B0) 25 are outputs of DATALAT3 10, DATALAT2 11,DATALAT1 12, DATALAT0 13, respectively, and have a latched logicalrelationship to the lines (DIN3) 14, (DIN2) 15, (DIN1) 16, (DIN0) 17,respectively. During page read cycle lines B3 22, B2 23, B1 24, B0 25are output of DATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13respectively and represent the 4 bits read out of the cell. Depending onthe status of lines (B3) 22, (B2) 23, (B1) 24, and (B0) 25, theREFERENCE MULTIPLEXER 26 couples one of the lines VR0 through VR15 toone input of the VOLTAGE COMPARATOR 27. The output of the VOLTAGECOMPARATOR 27 connects to the input of the LATCH 28. Under the controlof ALGOCNTRL 164, the line ENLATCOMP 29 functions as a strobe signal toenable the LATCH 28 during a certain time to latch the output of theVOLTAGE COMPARATOR 27. Line RBYLATCOMP 30 resets the LATCH 28 atsuitable times under the control of ALGOCNTRL 164. The PROGRAM/READCONTROL 31 outputs lines COMPOR 32 and COMPORB 33. COMPOR 32 and COMPORB33 lines are connected together in a wire-OR manner for all YDRV 110,SYDRV 114, and RYDRV 112. The PROGRAM/PROGRAM INHIBIT SWITCH 34 puts thememory cell coupled to it indirectly through line BLIN 35 into a programor program inhibit mode under the control of PROGRAM/READ CONTROL 31.Line BLIN 35 goes to the PSEL 120 for YDRV 110 and to blocks 124, 122for SYDRV 114, RYDRV 112 respectively. The lines VR0 through VR15individually are coupled to the output of a VRBUFFER 750.

FIG. 27 shows the details of a LATCH 28 block, a PROGRAM/READ CONTROL 31block and a PROGRAM/PROGRAM INHIBIT 34 block. The VROUT line 55 couplesthe output of REFERENCE MULTIPLEXER 26 to the positive input of aVOLTAGE COMPARATOR 27. The line COMPOUT 58 couples the output of theVOLTAGE COMPARATOR 27 to the D input of a latch 59. ENLATCOMP 29 goes tothe EN input of the latch 59. ENLATCOMP 29 acts as a strobe signal forthe latch. When ENLATCOMP 29 is at logic high the latch 59 outputs thelogic level on D input to the Q output. QB is the inverted logic levelof Q. When ENLATCOMP 29 goes to logic low, the latch 59 latches thelogic level on D input. RBYLATCOMP 30 goes to the reset R input of thelatch 59. When RBYLATCOMP 30 is logic low latch 59 is reset, whereby Qis at logic low and QB is at logic high. Line COMLATQ 40 couples the Qoutput of the latch 59 to the gate of a NMOS transistor N1 43. LineCOMLATQB 41 couples the QB output of the latch 59 to the gate of a NMOStransistor N2 44. Line COMLATQ 40 also couples to the data latchesDATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13. COMLATQ 40 alsocouples to one input of a 2 input NAND gate NAND 49. The other input ofthe NAND 49 is coupled to the signal READ2B. READ2B is at logic highduring page programming cycle and at logic low during page read cycle.The line NDO 52 couples the output of NAND 49 to the input of aninverter INV 48 and also to the gate inputs of PMOS transistor P1 45 andNMOS transistor N3 47. The line INVO 53 couples the output of INV 48 tothe gate of a PMOS transistor P2 46. Line BLIN 35 connects to oneterminal of each of P1 45, N3 47 and P2 46. BLIN 35 also connects to thenegative input of VOLTAGE COMPARATOR 27. The other terminal of P1 45 isconnected to inhibit voltage input VIH 57. Line N4D 54 connects theother terminals of N3 47 and P2 46 to one terminal of NMOS transistor N450. Line N5D 60 connects the other terminal of N4 50 to one terminal ofNMOS transistor N5 51. The other terminal of N5 51 is connected toground. The gates of N4 50 and N5 51 are connected to inputs VBIYDRVCAS56 and VBIYDRV 57 respectively. N4 50 and N5 51 form a current biascircuit whereby a constant current load is placed on the BLIN 35 whenINVO 53 is at logic low and NDO 52 is at logic high. N4 50 and N5 51together represent the predetermined bias current for the voltage modesensing as shown in FIG. 2C.

After the page program command and the address of the page to be programis issued, the data to be programmed is loaded in the data latchesDATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13 of each of the YDRVS10S, SYDRVS 114S or RYDRVS 112S. The REFERENCE MULTIPLEXER 26 thencouples one of the inputs VR0 through VR15 to its output VROUT 55.During a program verify cycle VREFD(0) through VREFD(15) are availableon the VR0 through VR15 lines respectively. VR0 through VR15 arecommonly coupled to REFERENCE MULTIPLEXER 26 of all the YDRV 110, SYDRV112, RYDRV 14. The REFYDRVS 116S have the data latches internally set.In this embodiment there are 16 REFYDRVS 116S. Each REFYDRVS 116S isused for a specific level. For example, the data latches of a REFYDRVS116S used for level 5 will be internally set to program level 5 intoreference cells coupled to it. VR0 through VR15 are commonly coupled toREFERENCE MULTIPLEXER 26 of all the REFYDRVS 116S. During a programverify cycle, VREFR(0) through VREFR(15) are respectively available atthe VR0 through VR15 lines of a REFYDRVS 116S. Depending on the outputB3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11, DATALAT112, DATALAT0 13 within each YDRVS 110S, SYDRVS 114S, SYDRVS 112S onespecific voltage VREFD(0) through VREFD(15) is output to the input ofthe VOLTAGE COMPARATOR 27. Depending on the output B3, B2, B1, B0 of thedata latches DATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13 withineach REFYDRV 116 one specific voltage VREFR(0) through VREFR(15) isoutput to the input of the VOLTAGE COMPARATOR 27.

The latch 59 within each REFYDRVS 116S, YDRVS 110S, SYDRVS 114S andRYDRVS 112S are all reset by pulsing line RBYLATCOMP 30. RBYLATCOMP 30is commonly connected to the reset input of the latch 59 within eachREFYDRVS 116S, YDRVS 110S, SYDRVS 114S, and RYDRVS 112S. After latch 59is reset, COMLATQ 40 is at logic low. The NAND 49 then outputs logichigh to line NDO 52. Output of INV 48 then is at logic low on line INVO53. With NDO 52 at logic high and INVO 53 at logic low transistors N3 47and P 246 couple BLIN 35 to N4 50. P1 45 de-couples the inhibit voltageVIH 57 from BLIN 35. The memory cell is placed in the voltage read modeand the cell read back voltage VCELLR(L) or VCELLD(L) is available onBLIN 35. At this point, the VOLTAGE COMPARATOR 27 compares the voltagesat its inputs. If voltage on BLIN 35 is higher then voltage on VROUT 55the output COMPOUT 58 is low, otherwise it is high. At this time apositive going strobe pulse is applied to the ENLATCOMP 29 common to allthe latches 59 in REFYDRVS 116S, YDRVS 110S, SYDRVS 114S and RYDRVS112S, to latch the status of line COMPOUT 58. If COMPOUT 58 is low, thenthe COMLATQ 40 remains at logic low.

If COMPOUT 58 is high, then the COMLATQ 40 switches to logic high. Ifduring an iteration of verify-program cycles any one of the latches 59latches a logic high on COMLATQ 40, called a program inhibit state, thenfor that specific REFYDRVS 116S, YDRVS 110S, SYDRVS 114S or RYDRVS 112S,the line NDO 52 is at low and the line INVO 53 is at logic high. Withlatch 59 in a program inhibit state, BLIN 35 is de-coupled from N4D 54and there is no current load, whereas, BLIN 35 is coupled to the inhibitvoltage VIH 57 through P1 45. With latch 59 in the program inhibitstate, further programming pulses do not cause programming.

The line COMPOR 32 is connected in a wire-OR fashion to all the COMPOR32 lines of each REFYDRVS 116S, YDRVS 110S, SYDRVS 114S or RYDRVS 112S.There is a pull up load coupling the COMPOR 32 line to the power supply.Similarly, the line COMPORB 33 is connected in a wire-OR fashion to allthe COMPORB 33 lines of each REFYDRVS 116S, DRVS 110S, SYDRVS 114S orRYDRVS 112S. There is a pull up load coupling the COMPORB 33 line to thepower supply. The COMPORB line 33 goes high whenever all the latches 59have reached the program inhibit mode. When the Program PulseCount=MAXPC is reached, the ALGOCNTRL 164 latches the status of COMPORBline 33 in a status latch in block INPUT LOGIC 160. The status latch canbe read at one of the IO[0:7] 1001 lines by the external host. IfALGOCNTRL 164 latches a logic low in the status latch in block INPUTLOGIC 160 then a program fail condition is reached and the ALGOCNTRL 164goes out of the page programming cycle.

If at the end of any verify-program iteration, the COMPOR 32 line goeshigh, the ALGOCNTRL 164 sequences to the margin verify mode. All latches59 are reset. All cells are placed in the voltage read mode by READB 52at logic low. At this time inhibit voltage is de-coupled from BLIN 35and current bias transistor N4 50 is coupled to BLIN 35. Cell voltagesVCELLR(L) or VCELLD(L) are respectively available on BLIN 35 of aREFYDRVS 116S or BLIN 35 of YDRVS 110S, SYDRVS 114S, or RYDRVS 112S.During program margin verify the voltages read back from the data cellsare checked for adequate margin from voltages read back from referencecells for each programmed level L. In the Upper Program Margin Verifymode, voltages UMV(0) through UMV(15) are placed on the VR0 throughVR(15). Depending on the output B3, B2, B1, B0 of the data latchesDATALAT3 10, DATALAT2 11, DATALAT1 12, DATALAT0 13 within each YDRVS110S, SYDRVS 114S, RYDRVS 112S one specific voltage UMV(O) throughUMV(15) is output to the input VROUT 55 of the VOLTAGE COMPARATOR 27. Atthis time the VOLTAGE COMPARATOR 27 compares the voltages at its inputs.If voltage on BLIN 35 is higher then voltage on VROUT 55 the outputCOMPOUT 58 is low, otherwise it is high. At this time a positive goingstrobe pulse is applied to the ENLATCOMP 29 common to all the latches 59in YDRVS 110S, SYDRVS 114S and RYDRVS 112S, to latch the status of lineCOMPOUT 58. If COMPOUT 58 is low, then the COMLATQ 40 remains at logiclow. If COMPOUT 58 is high, then the COMLATQ 40 switches to logic high.At this time, if LGOCNTRL 164 latches a logic low in the status latch inINPUT LOGIC 160 block by looking at the status of the COMPORB 33 line,then a program fail condition is reached and the ALGOCNTRL 164 goes outof the page programming cycle. Otherwise, ALGOCNTRL 164 sequences to theLower Program Margin Verify mode.

In the Lower Program Margin Verify mode, all latches 59 are reset.Voltages LMV(0) through LMV(15) are placed on the VR0 through VR(15).Depending on the output B3, B2, B1, B0 of the data latches (DATALAT3)10, (DATALAT2) 11, (DATALAT1) 12, (DATALAT0) 13 within each YDRVS 110S,SYDRVS 114S, RYDRVS 112S one specific voltage LMV(0) through LMV(15) isoutput to the input VROUT 55 of the VOLTAGE COMPARATOR 27. At this timethe VOLTAGE COMPARATOR 27 compares the voltages at its inputs. Ifvoltage on BLIN 55 is higher then voltage on VROUT 55 the output COMPOUT58 is low, otherwise is high. At this time a positive going strobe pulseis applied to the ENLATCOMP 29 common to all the latches 59 in YDRVS110S, SYDRVS 114S and RYDRVS 112S, to latch the status on line COMPOUT58. If COMPOUT 58 is low, then the COMLATQ 40 remains at logic low. IfCOMPOUT 58 is high, then the COMLATQ 40 switches to logic high. At thistime, if ALGOCNTRL 164 latches a logic low in the status latch in INPUTLOGIC 160 block by looking at the status of the COMPOR line 32, then aprogram fail condition is reached and the ALGOCNTRL 164 goes out of thepage programming cycle.

During page read cycle, after page read command and the page address isissued, the reference and the data cells are placed in the voltage readmode. At this time all the B3[0:3] lines output 1111. VR0 through VR15have VCELLR(0) through. VCELLR(15). VCELLR(0) through VCELLR(15) are thevoltages read out of the reference cells of the page being read. Underthe control of the ALGOCNTRL 164 block 4 bits are sequentially read intothe data latches (DATALAT3) 10, (DATALAT2) 11, (DATALAT1) 12, (DATALAT0)13. For example, B3 is read by forcing the output of DATALAT3 to outputB3=0. At this time B[0:3]=1110. The REFERENCE MULTIPLEXER 26 thenoutputs VCELLR(7) on the VROUT 55 in each of the YDRVS 110S, SYDRVS 114Sand RYDRVS 112S. The output COMPOUT 58 of the VOLTAGE COMPARATOR 27 ishigh or low depending on whether voltage VCELLD(L) on the BLIN 35 islower or higher relative to voltage VCELLR(7) on line VROUT 55. IfCOMPOUT 58 is high then a logic high is latched into DATALAT3 10 andB3=0, otherwise logic low is latched and B3=1. Next, B2 is read byforcing the output of DATALAT2 11 to output B2=0. At this timeB[0:3]=110B3. B3 is the output of DATALAT3 10 from previous sequence.The REFERENCE MULTIPLEXER 26 then outputs VCELLR(L), depending on 110B3on the VROUT 55 line in each of the YDRVS 110S, SYDRVS 114S and RYDRVS112S. The output COMPOUT 58 of the VOLTAGE COMPARATOR 27 is high or lowdepending on whether voltage VCELLD(L) on the BLIN 35 is lower or higherrelative to voltage VRCELL(L) on line VROUT 55. If COMPOUT 58 is highthen a logic high is latched into DATALAT2 11 and B2=0, otherwise logiclow is latched and B2=1. In this manner, the next two sequences latchtwo bits into the DATALAT1 12 and DATALAT0 13.

After all 4 bit from the cell are latched into the DATALAT3 10, DATALAT211, DATALAT1 12, DATALAT0 13 for all the YDRVS 110S, SYDRVS 114S andRYDRVS 112S, the restore margins are checked. All latches 59 are reset.First the RESTORE LOW margin is checked. At this time, for each level 0through 15, MARGIN RESTORE LOW Voltage VRSTRL(0) through VRSTRL(15) isplaced at the VR0 through VR15 lines respectively. Depending on eachoutputs B3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11,DATALAT1 12, DATALAT0 13 within each YDRVS 110S, SYDRVS 114S and RYDRVS112S, the REFERENCE MULTIPLEXER 26 outputs one of VRSTRL(0) throughVRSTRL(15) on line VROUT 55 going into the positive input of the VOLTAGECOMPARATOR 27. ENLATCOMP 29 is strobed with the positive pulse to latchthe status of the COMPOUT 58 line. If data cell read out voltageVCELLD(L) on BLIN 35 line is higher than voltage VRSTRL(L) on VROUT 55line then COMLATQ 40 remains at logic low and COMLATQB 41 at logic high.Otherwise, COMLAT 40 is at logic high and COMLATQB 41 at logic low. Atthis time, if ALGOCNTRL 164 latches a logic low in the RESTORE LOW latchin INPUT LOGIC 160 block by looking at the status of the COMPORB line33, then a restore low flag condition is reached. Next, all latches 59are reset.

Next the RESTORE HIGH margin is checked. At this time, for each level 0through 15, MARGIN RESTORE HIGH Voltage VRSTRH(0) through VRSTRH(15) isplaced at the VR0 through VR15 lines respectively. Depending on eachoutputs B3, B2, B1, B0 of the data latches DATALAT3 10, DATALAT2 11,DATALAT1 12, DATALAT0 13 within each YDRVS 10S, SYDRVS 114S and RYDRVS112S, the REFERENCE MULTIPLEXER 26 outputs one of VRSTRH(0) throughVRSTRH(15) on line VROUT 55 going into the positive input of the VOLTAGECOMPARATOR 27. ENLATCOMP 29 is strobed with the positive pulse to latchthe status of the COMPOUT 58 line. If data cell read out voltageVCELLD(L) on BLIN 35 line is higher than voltage VRSTRH(L) on VROUT 55line then COMLATQ 40 remains at logic low and COMLATQB 41 at logic high.Otherwise, COMLAT 40 is at logic high and COMLATQB 41 at logic low. Atthis time, if ALGOCNTRL 164 latches a logic low in the RESTORE HIGHlatch in INPUT LOGIC 160 block by looking at the status of the COMPORline 32, then a restore high flag condition is reached.

At this time, 4 bits from every cell with the page being read arelatched into the respective data latches within each YDRVS 110S, SYDRVS114S and RYDRVS 112S. Next under the control of the READ CL0CK data issequentially read on IO[0:7]. If after READ CL0CK the RED_ADD_TRUE=Ycondition is true then the data is read from the addressed RYDRVS 112Sotherwise data is read from the addressed YDRVS 110S or SYDRVS 114S.

FIG. 19B shows various voltages generated and used in another embodimentof the current invention for program verifying, program marginverifying, read sensing and restore high or low margin verifying. Inthis embodiment the program margin verify voltage VREFR(L)−VRM(L) andVREFD(L)−DM(L) for a level L of a reference cell and a data cellrespectively, are generated by the block V&IREF 172 independent of thevoltages VCELLR(L) and VCELLD(L) programmed into the reference cell anddata cell respectively. The voltage VRM(L) for a level L of thereference cells can be unique for each level or the same for all levels.The voltage VDM(L) for a level L of the data cells can be unique foreach level or the same for all levels.

FIG. 22B shows the portion of the flow for the page programming cyclethat uses the voltages as shown in FIG. 19B. In the flow shown in FIG.22B, only one program margin verify comparison is made instead of two asshown in FIG. 22A. This has the advantage of reducing the total time forcompletion of a page programming cycle.

FIG. 22C shows an alternative embodiment of the flow shown in FIG. 22B.At the end of the programming, a BSERV operation is done to verify thatthe read operation is operational versus the data in. The BSERVoperation is a binary search read verification operation that issubstantially the same as described in FIGS. 23 and 24 with theadditional step of comparing resulting digital bits BR<3:0> from thebinary search with a stored digital bits B<3:0> from loading data in. Ifthe comparison is not true, the program flag is set to indicate programfailure. The operation further ensures that all cells are within anoperational range, for example not out of range due to programmingovershoot to the next levels.

The embodiment shown in FIGS. 19B and 22B can be used in combinationwith the embodiment shown in FIGS. 19A and 22A. As discussed in themultilevel reference system section above, the embodiment shown in FIGS.19B and 22B can be used when a selected page programs for the first timeafter block erase. For subsequent page programming cycles on the samepage, the embodiment shown in FIGS. 19A and 22A is advantageous sincethe VCELLR(L) values may shift between initial page programming andsubsequent page programming.

FIG. 28 is a block diagram illustrating a memory system 2800 for amultilevel memory.

The memory cell 2800 comprises a plurality of memory arrays 2801arranged in rows and columns of memory arrays 2801. Each memory array2801 comprises a plurality of memory subarrays 2802, a plurality oflocal sense amplifiers 2804, and a plurality of global sense amplifiers2806. In one embodiment, a local sense amplifier 2804 is disposedadjacent to a memory subarray 2802. In another embodiment, the localsense amplifier 2804 is shared between a plurality of memory subarrays2802. The local sense amplifier 2804 reads the contents of the memorycells with the corresponding memory subarray 2802. The memory subarrays2802 are arranged in rows and columns. The local sense amplifiers 2804coupled to a column of memory subarrays 2802 are coupled to a globalsense amplifier 2806. The memory cells may include redundant cells,reference cells or spare cells.

FIG. 29A is a block diagram illustrating an inverter mode sensingcircuit 2900.

The inverter mode sensing circuit 2900 comprises a PMOS transistor 2902,a plurality of NMOS transistors 2904 and 2906, a feedback circuit 2908,a plurality of memory cells 2910, a comparator 2912. For clarity, onlyone memory cell 2910 and one NMOS transistor 2906 are shown for asubarray, but the subarray comprises a plurality of memory cells 2910arranged in columns. Each column has a corresponding NMOS transistor2906 or a plurality of NMOS transistors 2906 arranged in series. Onlyone column with one memory cell 2910 is shown.

The comparator 2912 determines the voltage of the memory cell bycomparing the cell voltage (VCELL) 2914 to a reference voltage (VREF)2916 in a manner described above. The PMOS transistor 2902, the NMOStransistors 2904 and 2906 and the memory cells 2910 are coupled inseries between the supply voltage and ground. The selected memory cell2910 is read by applying a control gate reference voltage (VCGRD) 2917on the control gate of the memory cell 2910. The column of memory cells2910 and an associated bitline has a capacitance 2918 that slows thesensing of the memory cells 2910. The NMOS transistor 2906 functions asa switch to couple the column of memory cells 2910 to the sensingportion of the circuit. The feedback circuit 2908 controls biasing ofthe NMOS transistor 2904 to stabilize the cell voltage 2914. The drainof the diode connected PMOS transistor 2902 is coupled to the cellvoltage 2914. Inverter mode sensing may also be referred to as currentmode sensing or common source sensing.

FIG. 29B is a block diagram illustrating a voltage mode sensing circuit2950.

The voltage sensing circuit 2950 is similar to the inverter mode sensingcircuit 2900 except that a current source 2952 replaces the PMOStransistor 2902 and is coupled to ground, the memory cell 2910 iscoupled to a reference bias, and the NMOS transistor 2904 and thefeedback circuit 2908 are omitted. The voltage mode sensing may also bereferred to as source follower sensing.

FIG. 30 is a block diagram illustrating a wide range, high speed voltagemode sensing circuit 3000.

The memory array 2800 includes a plurality of voltage mode sensingcircuits 3000. The voltage mode sensing circuit 3000 comprises a PMOStransistor 3002, a plurality of NMOS transistors 3004, 3006, 3007, afeedback circuit 3008, a plurality of memory cells 3010, a currentsource (IRCELL) 3011, and a comparator 3012. For clarity, only onememory cell 3010, one NMOS transistor 3006, and one NMOS transistor 3007are shown for a subarray, but the subarray comprises a plurality ofmemory cells 3010 arranged in columns. Each column has a correspondingNMOS transistor 3006. Only one column with one memory cell 3010 isshown. Possible decoding circuitry between the current source 3011 andthe memory cell 3010 and between the current source 3011 and the NMOStransistor 3007 is not shown.

The comparator 3012 determines the voltage of the memory cell bycomparing a cell voltage (VCELL) 3014 to a reference voltage (VREF) 3016in a manner described above. The PMOS transistor 3002, the NMOStransistors 3004, 3006 and 3007 are coupled in series between the supplyvoltage and ground. The current source 3011 is coupled between the gateof the NMOS transistor 3002 and ground. The memory cell 3010 is coupledbetween a reference voltage (VCLRD) and the common node formed of thecurrent source 3011 and the gate of the NMOS transistor 3007.

The selected memory cell 3010 is read by applying a control gatereference voltage (VCGRD) 3017 on the control gate of the memory cell3010. The biasing of the gate of the NMOS transistor 3007 by the currentsource 3011 and the memory cell 3010 controls the voltage on the bitline.

The NMOS transistor 3006 functions as a switch to couple the column ofNMOS transistors 3007 and the associated memory cells 3010 to thesensing portion of the circuit. The feedback circuit 3008 controlsbiasing of the NMOS transistor 3004 to stabilize the cell voltage 3014.The drain of the diode connected PMOS transistor 3002 is coupled to thecell voltage 3014.

FIG. 31 is a block diagram illustrating a voltage mode sensing circuit3100.

The voltage mode sensing circuit 3100 comprises a plurality of memorysubarrays 3150, a plurality of local sense amplifiers 3152, and aplurality of global sense amplifiers 3154. The local sense amplifier3152 includes a local source follower stage. The global sense amplifier3154 includes a common source stage.

The memory array 3150 includes columns of memory cells 3110 coupled tofirst bitlines 3151.

Each local sense amplifier 3152 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3152 is disposed adjacent thememory subarray 3150. The local sense amplifier 3152 includes aselection circuit 3153 that couples a selected bitline 3151 to a bitline3155. In one embodiment, the selection circuit 3153 comprisestransistors. The local sense amplifier 3152 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3154.

The local sense amplifier 3152 comprises an NMOS transistor 3107 coupledbetween the bitline 3155 and ground, and includes a gate coupled to thebitline 3151. A current source 3111 is coupled between the gate of theNMOS transistor 3107 and ground.

The global sense amplifier 3154 comprises a comparator 3112, a PMOStransistor 3102 and a selection circuit 3158. The selection circuit 3158couples the selected one of the bitlines 3155 to a common node formed ofa voltage cell input 3114 of the comparator 3112 and the drain of thediode connected PMOS transistor 3102. A reference voltage 3116 isapplied to the second input of the comparator 3112.

The local sense amplifier 3152 provides a larger voltage range by usingoptimally low current bias. The global sense amplifier 3154 includes acommon source stage with a PMOS transistor 3114 as a load, and buffersthe column capacitance.

The voltage mode sensing circuit 3100 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3172,and a plurality of global sense amplifiers 3174. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3172 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3172 are similar to the local sense amplifiers 3152. The global senseamplifiers 3174 detect and amplify the voltage from the local senseamplifiers 3172.

The global sense amplifier 3174 comprises a comparator 3173, a PMOStransistor 3174 and a selection circuit 3178, which are arranged insimilar manner as the comparator 3112, the PMOS transistor 3102 and theselection circuit 3158 of the global sense amplifier 3154, except thecomparator 3173 is configured as a buffer. The comparator 3173 serves asa comparator in sensing the reference cells and serves as a buffer fordriving the reference level.

FIG. 32 is a block diagram illustrating a voltage mode sensing circuit3200.

The voltage mode sensing circuit 3200 includes like elements as thevoltage mode sensing circuit 3100 (FIG. 31) and are given like referencenumbers. The voltage mode sensing circuit 3200 comprises a memory array3150, a plurality of local sense amplifiers 3252 and a plurality ofglobal sense amplifiers 3254. The local sense amplifier 3252 includes alocal source follower stage and includes a PMOS source follower as partof the global sense amplifier. The global sense amplifier 3254 includesa source follower stage.

Each local sense amplifier 3252 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3252 is disposed adjacent thememory subarray 3150. The local sense amplifier 3252 includes aselection circuit 3253 that couples a selected bitline 3151 to a bitline3255. In one embodiment, the selection circuit 3253 comprisestransistors. The local sense amplifier 3252 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3254.

The local sense amplifier 3252 comprises a PMOS transistor 3207 coupledbetween the bitline 3255 and ground, and includes a gate coupled to thebitline 3151. A current source 3211 is coupled between the gate of thePMOS transistor 3207 and ground. The local sense amplifier 3252 providesa maximum voltage range by using low current bias.

The global sense amplifier 3254 comprises a comparator 3212, a currentsource 3202 and a selection circuit 3258. The current source 3202couples the supply voltage to the cell voltage terminal 3214 of thecomparator 3212 to ground. The selection circuit 3258 couples theselected one of the bitlines 3255 to a common node formed of a voltagecell input 3214 of the comparator 3212 and the current source 3202. Areference voltage 3216 is applied to the second input of the comparator3212.

The global sense amplifier 3254 buffers the column capacitance.

The voltage mode sensing circuit 3200 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3282,and a plurality of global sense amplifiers 3274. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3282 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3282 are similar to the local sense amplifiers 3252. The global senseamplifiers 3274 detect and amplify the voltage from the local senseamplifiers 3282.

The global sense amplifier 3274 comprises a comparator 3292, a currentsource 3272 and a selection circuit 3278, which are arranged in similarmanner as the comparator 3212, the current source 3202 and the selectioncircuit 3258 of the global sense amplifier 3254, except the comparator3292 is configured as a buffer. The comparator 3292 serves as acomparator in sensing the reference cells and serves as a buffer fordriving the reference level.

FIG. 33 is a block diagram illustrating voltage mode sensing circuit3300.

The voltage mode sensing circuit 3300 includes like elements as thevoltage mode sensing circuit 3200 (FIG. 32) and are given like referencenumbers. The voltage mode sensing circuit 3300 comprises a memory array3150, a plurality of local sense amplifiers 3352 and a plurality ofglobal sense amplifiers 3354. The local sense amplifier 3352 includes alocal source follower stage and includes an NMOS source follower as partof the global sense amplifier. The global sense amplifier 3354 includesa source follower stage.

Each local sense amplifier 3352 is coupled to a memory subarray 3150. Inone embodiment, the local sense amplifier 3352 is disposed adjacent thememory subarray 3150. The local sense amplifier 3352 includes aselection circuit 3253 that couples a selected bitline 3151 to a bitline3355. In one embodiment, the selection circuit 3253 comprisestransistor. The local sense amplifier 3252 senses the selected memorycell on the bitline 3151 and provides a voltage to a global senseamplifier 3254.

The local sense amplifier 3352 comprises an NMOS transistor 3307 coupledbetween the bitline 3355 and a supply voltage terminal, and includes agate coupled to the bitline 3151. A current source 3311 is coupledbetween the gate of the NMOS transistor 3307 and ground. The local senseamplifier 3252 provides a maximum voltage range by using low currentbias.

The global sense amplifier 3354 comprises a comparator 3312, a currentsource 3302 and a selection circuit 3358. The current source 3302couples the voltage terminal 3314 of the comparator 3312 to a groundterminal. The selection circuit 3358 couples the selected one of thebitlines 3355 to a common node formed of a voltage cell input 3314 ofthe comparator 3312 and the current source 3302. A reference voltage3316 is applied to the second input of the comparator 3312. The globalsense amplifier 3354 is selectively coupled to the bitline to comparethe cell voltage to a reference voltage 3316. The global sense amplifier3354 buffers the column capacitance.

The voltage mode sensing circuit 3300 further comprises a plurality ofreference subarrays 3170, a plurality of local sense amplifiers 3382,and a plurality of global sense amplifiers 3374. The reference subarrays3170 comprise a plurality of reference cells for storing referencesignals. In one embodiment, the reference subarrays 3170 are similar tothe memory subarrays 3150. The local sense amplifiers 3382 read thereference subarrays 3170. In one embodiment, the local sense amplifiers3382 are similar to the local sense amplifiers 3352. The global senseamplifiers 3374 detect and amplify the voltage from the local senseamplifiers 3382.

The global sense amplifier 3374 comprises a comparator 3392, a currentsource 3372 and a selection circuit 3378, which are arranged in similarmanner as the comparator 3312, the current source 3302 and the selectioncircuit 3358 of the global sense amplifier 3354, except the comparator3392 is configured as a buffer. The comparator 3392 serves as acomparator in sensing the reference cells and serves as a buffer fordriving the reference level.

In another embodiment, the local sense amplifier is a common sourceamplifier, and the global sense amplifiers are NMOS source followerstages or PMOS source follower stages.

In another embodiment, the local sense amplifier is a common sourceamplifier, and the global sense amplifiers are common source amplifiers.

FIG. 34 is a block diagram illustrating a global sense amplifier 3400having an auto zeroing function.

The comparators 3012, 3112, 3212, and 3312 of FIGS. 30-33 may be theglobal sense amplifier 3400.

The sense amplifier 3400 comprises an operational amplifier 3402, a pairof capacitors 3404 and 3405, and a plurality of switches 3406 and 3407.

The capacitors 3404 and 3405 couples respective inputs 3408 and 3410 ofthe operational amplifier 3402 to the switch 3406.

In response to an auto zero (AZ) command 3416, the switches 3407selectively couples an output 3412 of the operational amplifier 3402 tothe input 3408 to equalize the voltages on the output 3412 and input3408, and selectively couples an output 3414 of the operationalamplifier 3402 to the input 3410 to equalize the output 3414 and theinput 3410. In the auto zero mode, the voltage on A terminals of thecapacitors 3404 and 3405 are set equal to the reference voltage (VREF)3418, and the B terminals of the capacitors 3404 and 3405 are equalizedto the complementary outputs of the operational amplifier 3402. Theswitch 3406 is switched by an evaluation (EVA) command 3422 to connectthe cell voltage (VCELL) 3420 to the other end of the capacitor 3405 forcomparison from the operational amplifier 3402.

The switch 3406 selectively applies the reference voltage (VREF) 3418 tothe capacitor 3404 in response to the evaluation (EVA) command 3422. Theswitch 3406 also selectively applies either the reference voltage (VREF)3418 or a cell voltage (VCELL) 3420 to the capacitor 3405 in response tothe evaluation (EVA) command 3422. The evaluation command 3422 equalizesthe signals on terminals 3404A and 3505A of the capacitors 3404 and3405.

In an alternate embodiment, the nodes 3404B and 3405B of the capacitors3404 and 3405 are reset to a fixed bias voltage. In another embodiment,the nodes 3404B and 3405B of the capacitors 3404 and 3405 are shortedtogether.

By using a capacitor for sensing, the input common load range to theoperational amplifier (or comparator) is substantially constant andindependent of the memory cell voltage or current.

FIG. 35 is a block diagram illustrating an auto zero sense amplifier3500.

The autozero sense amplifier 3500 comprises a plurality of PMOStransistors 3502 and 3504, a plurality of NZ NMOS transistors 3506 and3507, a plurality of NMOS transistors 3508 through 3516, a plurality ofcapacitors 3518 and 3519 and a plurality of transfer gates 3522 through3528.

The PMOS transistors 3502 and 3504 and the NMOS transistors 3508, 3509and 3513 and the NZ NMOS transistor 3507 are arranged as a differentialpair. The NMOS transistors 3508 and 3509 provide the differential inputpair. The NZ NMOS transistor 3507 and the NMOS transistor 3513 providebias for the NMOS transistor 3508 and 3509. The PMOS transistors 3502and 3504 are coupled for cross-coupled loading. The PMOS transistor 3502is coupled between the supply voltage and an output terminal 3530. Abias voltage 3529 is applied to the gates of the NZ NMOS transistors3506 and 3507 and the NMOS transistors 3513 and 3514.

The NMOS transistors 3510 and 3511 provide an NMOS coupled internallatch, which is active while the differential input pair is on. Thedrain of the NMOS transistor 3510 is coupled to the drain of the NMOStransistor 3509 and the gate of the NMOS transistor 3511. The drain ofthe NMOS transistor 3511 is coupled to a common node formed of the drainof the NMOS transistor 3508 and gate of the NMOS transistor 3510. The NZNMOS transistor 3506 and the NMOS transistor 3514 provide bias for theNMOS transistors 3510 and 3511 and are coupled between the common nodeformed of the sources of the NMOS transistors 3510 and 3511, and ground.

The transfer gate 3522 couples the drains of the PMOS transistors 3502and 3504 and the output 3530 to each other for equalization and quickrecovery for the next comparison in response to a release signal 3531and an inverted release signal 3532.

The capacitor 3519 couples the gate of the NMOS transistor 3509 to firstterminals of the transfer gates 3525 and 3526 which include a secondterminal coupled to a reference voltage 3534. The capacitor 3518 couplesthe gate of the NMOS transistor 3508 into first terminals of thetransfer gates 3527 and 3528, which have second terminals coupled to thereference voltage 3534 and a cell voltage 3535, respectively. Thetransfer gates 3525 and 3527 are controlled by a auto zero signal 3537and an inverted auto zero signal 3538. The transfer gates 3526 and 3528are controlled by evaluation signals 3539 and 3540.

The transfer gates 3523 and 3524 couple the drains of the PMOStransistors 3504 and 3502, respectively, to the gates of the NMOStransistors 3509 and 3508, respectively, in response to the auto zerosignal 3537 and inverted auto zero signal 3538. The NMOS transistors3512 and 3516 couple the gates of the NMOS transistors 3509 and 3508,respectively, to ground in response to a strobe signal 3542 to pull downthe transistors 3509 and 3508 to turn off the differential pair. TheNMOS transistor 3515 couples the sources of the NMOS transistors 3510and 3511 to the ground in response to the strobe signal 3542 for fulllevel latching.

The array architectures described herein may enable multilevel paralleloperation.

A pipelined read operation may be as follows. A first row is selected ina selected subarray, such as subarray 2802 or subarray 3150/3170, andthe content of selected memory cells are coupled to the local bitlineand to the global bitlines while a second row in another subarray 2802or 3150/3170 is selected and the content of the selected memory cellsare coupled to the local bitlines but not yet coupled to the globalbitlines. After the read operation completes processing the data of thefirst row, the data of the second row is enabled to couple to the globalbitlines to continue the read operation, and a third row in a differentsubarray 2802 or 3150/3170 is selected to enable the content of theselected memory cells to couple to the local bitlines but not yet to theglobal bitlines. This cycle continues until all desired data are readout. This, for example, enables continuous read of multilevel memorycells.

In another embodiment, pipelined read operation is performed byoperating on memory cells in a row in an array, such as memory array2801, while another row in another memory array 2801 is selected toenable the contents of the memory cells to be ready.

A read-while-read operation may be as follows. A read operation operateson both arrays, such as memory array 2801 (or memory subarrays 2802 or3150), simultaneously and the data are available from both arrayspossibly at the same time. In this case, for example, data latches areused to latch the data from both arrays. In another embodiment, two setsof data lines may be used to transfer the data from both arrays to anon-chip controller.

A read/write-while-write/read operation may be as follows. Similarlywhile one operation, e.g., read, is executed on an array, such assubarray 2802 or array 2801 or subarrays 3150/3170, another operation isexecuted, e.g., write, on another array such as subarray 2802 or 2801 orsubarray 3150/3170. This is possible because control circuits associatedwith decoding and sensing and/or writing may be embedded for each array.

A read/erase-while-erase/read may be as follows. Similarly while oneoperation, e.g., read, is executed on an array, such as subarray 2802 or2801 or subarray 3150/3170, another operation is executed, e.g., erase,on another array such as subarray 2802 or 2801 or subarray 3150/3170.This is possible because each array may have its own decoders andembedded control circuits associated with sensing.

An erase-while-erase operation may be as follows. Similarly while oneerase operation is executed on an array, such as subarray 2802 or 2801or subarray 3150/3170, another erase operation is executed on anotherarray, such as subarray 2802 or 2801 or subarray 3150/3170. This ispossible because each array may have its own decoders.

A write/erase-while-erase/write operation may be as follows. Similarlywhile one operation, e.g., write, is executed on an array, such assubarray 2802 or array 2801 or subarrays 3150/3170, another operation isexecuted, e.g., erase, on another array such as subarray 2802 or 2801 orsubarray 3150/3170. This is possible because each array may have its owndecoders and embedded control circuits associated with sensing and/orwriting.

A write-while-write operation may be as follows. Similarly while onewrite operation is executed on an array, such as subarray 2802 or 2801or subarray 3150/3170, another write operation is executed on anotherarray, such as subarray 2802 or 2801 or subarray 3150/3170. This ispossible because each array may have its own decoders and embeddedcontrol circuits associated with sensing and/or writing.

In the foregoing description of various method and apparatus, it wasreferring to various specific embodiments. However it should be obviousto the one conversant in the art, various alternatives, modifications,and changes may be possible without departing from the spirit and thescope of the invention which is defined by the metes and bounds of theappended claims.

1.-52. (canceled)
 53. A data storage system comprising: a plurality ofreference memory subarrays, each reference memory subarray comprising aplurality of reference memory cells, each memory cell being configurableto store one of a plurality of reference signal levels; and a pluralityof reference sense amplifiers, each reference sense amplifier beingselectively coupled to a corresponding subarray; a capacitor configuredto capacitively sense content of a memory cell.
 54. (canceled)
 55. Adata storage system comprising: a plurality of reference memorysubarrays, each reference memory subarray comprising a plurality ofreference memory cells, each reference memory cell being configurable tostore one of a plurality of reference signal levels; and a plurality ofreference sense amplifiers, each sense reference amplifier beingselectively coupled to a reference memory subarray to sense with offsetautozeroing content of a reference memory cell. 56.-126. (canceled) 127.The data storage system of claim 53 wherein each reference senseamplifier is selectively coupled to a plurality of reference memorysubarrays to capacitively sense an output of a memory cell.
 128. Thedata storage system of claim 53 further comprising a reference arrayoperatively coupled to the reference memory subarrays and configurableto provide stored reference signals used for programming and readingselected reference memory cells, the stored reference signalscorresponding to detected reference signals.
 129. The data storagesystem of claim 128 wherein each reference sense amplifier isselectively coupled to a plurality of reference memory subarrays tocapacitively sense an output of a memory cell.
 130. The data storagesystem of claim 53 wherein the reference sense amplifier capacitivelydetects a signal on a bitline and a reference signal and generates anoutput signal indicative of the comparison between said detected signalin a selected reference cell and said detected reference signal. 131.The data storage system of claim 130 wherein each reference senseamplifier is selectively coupled to a plurality of reference memorysubarrays to capacitively sense an output of a memory cell.
 132. Thedata storage system of claim 130 further comprising a reference arrayoperatively coupled to the reference memory subarrays and configurableto provide stored reference signals used for programming and readingselected reference memory cells, the stored reference signalscorresponding to detected reference signals.
 133. The data storagesystem of claim 132 wherein each reference sense amplifier isselectively coupled to a plurality of reference memory subarrays tocapacitively sense an output of a memory cell.
 134. The data storagesystem of claim 53 wherein each reference sense amplifier beingselectively coupled to a group of sense amplifiers to capacitively sensean output associated with said group of sense amplifiers.
 135. The datastorage system of claim 53 wherein each reference sense amplifier beingcoupled to a group of bitlines to capacitively detect a signal on abitline and to a reference array to capacitively detect a referencesignal, and to generate an output signal indicative of the comparisonbetween said detected signal in a selected memory cell and said detectedreference signal.
 136. The data storage system of claim 55 wherein eachreference sense amplifier autozeros an input and output in order to zeroout offset of reference sense circuitry.
 137. The data storage system ofclaim 55 wherein at least one reference sense amplifier includes acircuit for coupling inputs of the at least one reference senseamplifier to an output signal in response to an autozero signal. 138.The data storage system of claim 137 wherein each reference senseamplifier autozeros an input and output in order to zero out offset ofreference sense circuitry.
 139. The data storage system of claim 55further comprising a reference array operatively coupled to thereference memory subarrays and configurable to provide stored referencesignals used for programming and reading selected reference memorycells, the stored reference signals corresponding to detected referencesignals.
 140. The data storage system of claim 139 wherein eachreference sense amplifier autozeros an input and output in order to zeroout offset of reference sense circuitry.
 141. The data storage system ofclaim 139 wherein at least one reference sense amplifier includes acircuit for coupling inputs of the at least one reference senseamplifier to an output signal in response to an autozero signal. 142.The data storage system of claim 141 wherein each reference senseamplifier auto zeros an input and output in order to zero out offset ofreference sense circuitry.
 143. The data storage system of claim 55further comprising a circuit to set an input and an output of referencecircuitry in response to an autozero signal.
 144. The data storagesystem of claim 143 wherein reference circuitry autozeros an input andan output in order to zero out an offset of the reference circuitry.